Blogshttp://e2e.ti.com/blogs_/Welcome to the Blogs section of the TI E2E Community! Ask questions, share knowledge, explore ideas, and help solve problems with fellow engineers on TI’s Engineer-to-Engineer (E2E) Communityen-USZimbra Community 8Blog Post: How to make precision measurements on a nanopower budget, part 2: current sensing using nanopower op ampshttp://e2e.ti.com/blogs_/b/analogwire/archive/2018/02/08/how-to-make-precision-measurements-on-a-nanopower-budget-part-2-current-sensing-using-nanopower-op-ampsThu, 08 Feb 2018 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:ee3b8d9d-78d7-4431-a4fe-2f706eafecf6Gen VansteegIn the first installment of this two-part series, I discussed the mechanism offset voltage (V OS ) and offset voltage drift (TCV OS ) in DC gain, and how to select a nanopower operational amplifier (op amp) with the right level of precision to minimize error in the signal path of an amplified low-frequency signal. In this second installment, I’ll review some basics of current sensing and show ways in which you can use an op amp to help minimize system power consumption while still providing accurate readings. Current sensing Designers perform current sensing for system protection and monitoring by placing a very small “shunt” resistor in series with the load in question, across which there’s a current-sense amp or an op amp. Purpose-built current-sense amps are exceptionally good at sensing current, but in cases where power consumption is most important, a precision nanopower op amp is a good alternative. There are two positions in which you can place the shunt resistor with respect to the load: between the load and the supply (Figure 1), or between the load and ground (Figure 2). Figure 1: High-side current sensing Figure 2: Low-side current sensing In both cases, to sense current with a known resistor value, the op amp measures the voltage across the terminals of the shunt resistor. Using Ohm’s law (Equation 1), you can determine the current consumption: where V is the voltage, I is the current and R is the resistance. Choose the shunt resistor and op amp so that they will have minimal impact on the circuit’s behavior. In selecting a resistor, there are two factors driving the need for a low-value resistor: Keeping the voltage drop across the resistor as small as possible so that the load’s negative terminal is as close to ground as possible in low-side sensing, or as close to the supply as possible in high-side sensing. Keeping power dissipation low. Equation 2 shows that, since you are measuring current and it is therefore the independent variable, the resistance should be as small as feasible: A point of clarification: Since you are measuring current rather than trying to minimize current (as I did in the first installment of this series), you need to minimize the value of the resistor to also minimize power dissipation – this is the opposite thought process of managing power dissipation in DC gain configurations. Ultra-low-power current-measurement techniques are useful in end equipment like power bank battery charging and monitoring, mobile phone battery charging and monitoring, and even to ensure the correct functioning of industrial Internet of Things applications. So just how low can you go when selecting resistor values? Simply put, the voltage drop across the resistor should be larger than the offset voltage of the op amp you’re using. Example Let’s say that you want to perform low-side differential current measurement (Figure 3) to ensure that there is neither a short nor an open connection in your system. For the purposes of simplicity, let’s choose easy numbers for this example and ignore parameters like resistor tolerance. Figure 3: Low-side differential current measurement The supply voltage is 3.3V. The system draws a maximum of 10mA when operating correctly, and you don’t want the effective ground that the load sees to be any higher than 100µV. The first thing you’ll realize is that your shunt resistor must exhibit a voltage drop (due to current) less than or equal to 100µV. If you use Equation 3 to determine the maximum shunt resistance: Your effective ground will be 100µV, as shown in Equation 4: You must choose an op amp that is able to detect changes in this voltage drop, which indicate a fault condition. Since the system is in a normal operating condition when the load current is within ±10% of its typical value of 10mA. Your op amp will then be able to detect the change in voltage across the sense resistor when the current changes by at least 10%. When there is a fault condition (such as an open circuit, a brownout from low current, or a short-circuit or brownout from high current), Equation 5 expresses the change in current (I Δ ) as: Equation 6 calculates the resulting change in voltage drop across V SHUNT : In this example, I would choose the LPV821 zero-drift nanopower amplifier. Its zero-drift technology enables a maximum offset voltage of only 10µV, allowing it to detect the fault condition. Zero-drift op amps are ideal for high accuracy (<100µV) measurements. Additionally, the LPV821 is also a nanopower amplifier, which means you can leave it always-on, and it will continue to deliver accurate current-sense measurements with very little impact to the system power budget. Thanks for reading this second installment of the “How to make precision measurements on a nanopower budget” series. We hope that this series provided some insight on the benefits of using nanopower zero-drift op amps in DC gain and low-side current-sensing applications. If you have any questions about precision measurements, log in and leave a comment, or visit the TI E2E™ Community Precision Amplifiers forum . Additional resources Download the LPV821 data sheet. For more configurations of op amps in current sensing applications or about designing on a nanopower budget, check out these TI TechNotes: “ Advantages of Using Nanopower, Zero Drift Amplifiers for Battery Voltage and Current Monitoring in Portable Applications .” “ Current Sensing in No-Neutral Light Switches .” “ Simplifying Measurements in Power Conscious Industrial Analytics Systems with Nanopower Op Amps .” Check out all of TI’s amplifier devices .opampportalcurrent sensing blogPrecision Op Amps (Vos<1mV)LPV821analog blogpampportalUltra-Low-Power Op Amps (IQ<=10µA)Blog Post: How to make precision measurements on a nanopower budget, part 1: DC gain in nanopower op ampshttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/12/06/how-to-make-precision-measurements-on-a-nanopower-budgetWed, 06 Dec 2017 11:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:44801342-a5f4-4a88-995e-320e31fd2293Gen VansteegHeightened accuracy and speed in an operational amplifier (op amp) has a direct relationship with the magnitude of its power consumption. Decreasing the current consumption decreases the gain bandwidth; conversely, decreasing the offset voltage increases the current consumption. Many such interactions between op amp electrical characteristics influence one another. With the increasing need for low power consumption in applications like wireless sensing nodes, the Internet of Things (IoT) and building automation , understanding these trade-offs has become vital to ensure optimal end-equipment performance with the lowest possible power consumption. In the first installment of this two-part blog post series, I’ll describe some of the power-to-performance trade-offs of DC gain in precision nanopower op amps. DC gain You probably remember from school the classic inverting (Figure 1) and noninverting (Figure 2) gain configurations of op amps. Figure 1: Inverting op amp Figure 2: Noninverting op amp These configurations resulted in inverting and noninverting op amp closed-loop gain equations, Equations 1 and 2, respectively: where A CL is the closed-loop gain, R F is the value of the feedback resistor and R 2 is the value of the resistor from the negative input terminal to signal (inverting) or ground (noninverting). These equations are a reminder that DC gain is based on resistor ratio, not resistor value. Additionally, the “power” law and Ohm’s law show the relationships between resistor value and power dissipation (Equation 3): where P is the power consumed by the resistor, V is the voltage drop across the resistor and I is the current through the resistor. For nanopower gain and voltage divider configurations, Equation 3 tells you that, in order to minimize power dissipation, you need to minimize the current consumption by the resistor. Equation 4 helps you understand the mechanism to do that: where R is the resistor value. Using these equations, you can see that you must choose large resistor values that provide both the gain you need while minimizing power dissipation (and therefore power consumption). If you don’t minimize current through the feedback path, you’ll lose the benefit of using nanopower op amps. Once you’ve determined what resistor values will meet your gain and power-consumption needs, you’ll need to consider some of the other op amp electrical characteristics that will affect the accuracy of signal conditioning. Summing several small systemic errors inherent in nonideal op amps will give you the total offset voltage. The electrical characteristic, V OS , is defined as a finite offset-voltage number between the op amp inputs, and describes these errors at a defined bias point. Please note that it does not describe these errors across all operating conditions. To do that, you must consider the gain error, bias current, voltage noise, common-mode rejection ratio (CMRR), power-supply rejection ratio (PSRR) and drift. Covering all of these parameters is beyond the scope of this post, but let’s look at V OS and drift – and their influence in nanopower applications – in a bit more detail. Real-world op amps exhibit V OS across their input terminals, which can sometimes be a problem in low-frequency (close to DC) precision signal-conditioning applications. In voltage gain configurations, the offset voltage will gain up along with the signal being conditioned, introducing measurement errors. In addition, the magnitude of V OS can change over both time and temperature (drift). Therefore, in low-frequency applications requiring fairly high-resolution measurements, it’s important to select a precise ( V OS ≤ 1mV) op amp with the lowest possible drift. Equation 5 calculates the worst-case V OS over temperature: Now that I’ve covered theory, including choosing large resistor values to create gain ratios and op amp accuracy for low-frequency applications, I’ll go over a practical example using two-lead electrochemical cells. For two-lead electrochemical cells which often emit very small signals of low frequency, and are used in diverse portable sensing applications like gas detection and blood glucose monitoring, choose a low-frequency (<10kHz) nanopower op amp. Using oxygen sensing (see Figure 3) as the specific application example, assume that the maximum concentration of the sensor outputs 10mV (converted from current to voltage by a manufacturer-specified load resistor, R L ) and the full-scale output of the op amp is 1V. Using Equation 2, you can see that A CL needs to be 100, or R F needs to be 100 times larger than R 2 . Choosing values of 100MΩ and 1MΩ, respectively, gives you a gain of 101, and these resistor values are large enough to limit current and minimize power consumption. To minimize offset error, the LPV821 zero-drift nanopower op amp is a good choice. Using Equation 5 and assuming an operating temperature range from 0°C to 100°C, the worst-case offset error introduced by this device will be: Another good choice is the LPV811 precision nanopower op amp. Using its data sheet to gather the necessary values plugged into Equation 5 gives you: (Note that the LPV811 data sheet does not specify a maximum offset voltage drift limit, so I am using the typical value here.) If you were to use a general-purpose nanopower op amp like the TLV8541 instead, those values would result in: (The TLV8541 data sheet also does not specify a maximum offset voltage drift limit, so I again used the typical value here.) As you can see, the LPV821 op amp is the best choice for this application. With 650nA of current consumption, the LPV821 can sense changes in the output of the oxygen sensor down to 18µV or lower, and introduces a maximum offset gain error of only 2.3mV. When you need both extreme precision and nanopower consumption, a zero-drift nanopower op amp will provide the best possible performance. Thanks for reading this first installment of the “How to make precision measurements on a nanopower budget” series. In the next installment, I’ll discuss how ultra-precise nanopower op amps can help in current-sensing applications. If you have any questions about precision measurements, log in and leave a comment, or visit the TI E2E™ Community Precision Amplifiers forum . Additional resources Download the LPV821 , LPV811 and TLV8541 data sheets. Read about designing on a nanopower budget in these TI TechNotes: “ Current Sensing in No-Neutral Light Switches.” “ Advantages of Using Nanopower Zero-Drift Amplifiers for Battery Voltage and Current Monitoring in Portable Applications .” “ Simplifying Measurements in Power-Conscious Industrial Analytics Systems with Nanopower Op Amps .” Check out all of TI’s amplifier devices .opampportalTLV8541Precision Op Amps (Vos<1mV)LPV821LPV811pampportalUltra-Low-Power Op Amps (IQ<=10µA)Blog Post: “Trust, but verify” SPICE model accuracy, part 6: voltage noise and current noisehttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/12/04/trust-but-verify-spice-model-accuracy-part-6-voltage-noise-and-current-noiseMon, 04 Dec 2017 17:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:d5da8e7a-3908-4819-aee9-849c8ae16b8aIan WilliamsPrevious installments of this blog post series discussed the need to verify SPICE model accuracy and how to measure common-mode rejection ratio (CMRR), offset voltage versus common-mode voltage (Vos vs. Vcm), slew rate (SR), open-loop output impedance (Zo), input offset voltage (Vos) and open-loop gain (Aol). In this sixth and final installment, I’ll cover operational amplifier (op amp) noise, including voltage noise and current noise. Noise is simply an unwanted signal, usually random in nature, that when combined with your desired signal results in an error. All op amps, as well as certain other circuit elements like resistors and diodes, generate some amount of intrinsic – or internal – noise. In analog circuits, it’s critical to confirm that the noise level is low enough to obtain a clear measurement of your desired output signal. Figure 1 shows an example of input voltage, ideal output voltage and output voltage with noise for a circuit with gain of 3V/V. Figure 1: Noise example With an accurate model, predicting the noise performance of an op amp circuit becomes quite straightforward. This is very appealing to most engineers, as calculating noise by hand can be cumbersome and difficult. Input voltage noise density The voltage noise of an op amp is usually given as input voltage noise density (e n ) in nanovolts per square root hertz (nV/√Hz), which quantifies how much noise voltage the op amp generates at its input pins for any given frequency. To measure e n , configure the op amp as a unity gain buffer with its noninverting input connected to an AC source Vin. Figure 2 shows the recommended test circuit. Figure 2: Input voltage noise density test circuit Let’s use this circuit to measure the e n of the OPA1692 , a low-noise amplifier from TI. Simply run a noise analysis over the desired frequency range and measure the noise level at node Vnoise with respect to Vin. In this case, the simulated e n matches perfectly with the data-sheet spec, shown in Figure 3. Figure 3: OPA1692 e n result Input current noise density Op amps also generate noise currents at their input pins, called input current noise density (i n ) and typically given in femtoamperes per square root hertz (fA/√Hz). You can measure this in a similar way to e n , but you will need to perform a simple trick. Some simulators have trouble measuring noise in terms of current, so a current-controlled voltage source converts the current flowing into the noninverting input pin into a voltage. Figure 4 shows the recommended test circuit. Figure 4: Input current noise density test circuit Let’s use this circuit to measure the i n of the OPA1692 . Run a noise analysis over the desired frequency range and measure the noise level at node Inoise with respect to Vin. Keep in mind that the resulting plot will have converted amps to volts due to current-controlled voltage source (CCVS1). Figure 5 shows the results after converting back to amperes. Figure 5: OPA1692 i n result Again, the noise characteristic matches the data-sheet curve extremely well. Total voltage noise While knowing the input-referred noise of an op amp is useful, it doesn’t paint a complete picture of your circuit’s overall noise performance. A combination of factors like closed-loop gain, bandwidth and the noise contributions of other circuit elements will affect the total amount of noise that appears at the circuit output. Thankfully, most simulators provide a way to measure this type of noise, called total noise or integrated noise, since it’s the integration of all noise sources over the circuit’s effective bandwidth. Figure 6 shows a more complex op amp circuit, with the OPA1692 configured for a noninverting gain of 10V/V and an additional resistor-capacitor (RC) filter at the output to limit the effective bandwidth to roughly 150kHz. Figure 6: OPA1692 total noise example circuit Run a total noise analysis over a wide frequency range (shown in Figure 7) and measure the noise level at node Vnoise in order to find the total root mean square (RMS) noise, which will appear at the circuit output. You are looking for the level at which the total noise curve flattens out to a constant value at high frequency. Figure 7: OPA1692 total noise result The test result shows that the total noise of the circuit in Figure 6 is equal to 21.15µVrms, or 126.9µVpp. This is what you would expect to measure if you probed the output of this circuit in the real world. However, keep in mind that the random nature of noise means that the actual noise level may be somewhat higher or lower than what you calculated or simulated. For a deeper discussion, watch the TI Precision Labs – Op Amps video series on noise . Thanks for reading this sixth and final installment of the “Trust, but verify” blog series! I hope you’ve found the information and techniques in this series useful in your pursuit of more accurate SPICE simulations. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Download the OPA1692 datasheet . Learn more from industry expert Art Kay who literally wrote the book on noise .opampportalPrecision Op Amps (Vos<1mV)OPA1692SPICE blogpampportalBlog Post: “Trust, but verify” SPICE model accuracy, part 5: input offset voltage and open-loop gainhttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/11/20/trust-but-verify-spice-model-accuracy-part-5-input-offset-voltage-and-open-loop-gainMon, 20 Nov 2017 17:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:176feb42-ba7d-4efe-824a-dffa5c8e8113Ian WilliamsPrevious installments of this blog post series discussed the need to verify SPICE model accuracy and how to measure common-mode rejection ratio (CMRR), offset voltage versus common-mode voltage (Vos vs. Vcm), slew rate (SR) and open-loop output impedance (Zo). In part 5, I’ll explain how to verify two of the most impactful specs of precision operational amplifiers (op amps): input offset voltage (Vos) and open-loop gain (Aol). Input offset voltage Vos is the difference in voltage between an op amp’s two input pins. Typical offset voltages range from millivolts down to nanovolts, depending on the device. Vos adds in series with any externally applied input voltage (Vin), and therefore can cause errors if Vos is significant compared to Vin. For this reason, op amps with low Vos are highly desirable for precision circuits with small input voltages. Figure 1 shows the application of a 1mV input voltage to an op amp with Vos equal to 0.1mV. Because Vos is 10% of Vin, the offset voltage contributes a 10% error in the overall circuit output. While this is a fairly extreme example, it shows the impact that Vos can have on op amp designs. Figure 1: Input offset voltage contribution to DC error To measure the Vos of an op amp, configure the op amp as a unity gain buffer with its noninverting input connected to mid supply (ground in split-supply circuits). Wire a differential voltage probe between the op amp input pins, and make sure to match the power-supply voltage and common-mode voltage conditions given in the op amp data sheet. Figure 2 shows the recommended test circuit. Figure 2: Input offset voltage test circuit Let’s use this circuit to measure the Vos of the OPA189 , a new zero-drift, low-noise amplifier from TI. Simply run a DC analysis and observe the voltage at probe Vos, as shown in Figure 3. Figure 3: OPA189 Vos result The measured input offset voltage is -400nV, or -0.4µV. This correlates exactly with the spec in the OPA189 data sheet . Open-loop gain An op amp’s open-loop gain is arguably its most important parameter, affecting nearly all aspects of linear or small-signal operation including gain bandwidth, stability, settling time and even input offset voltage. For this reason, it’s essential to confirm that your op amp SPICE model matches the behavior given in the device’s data sheet. Figure 4 shows the recommended test circuit. Figure 4: Open loop gain test circuit This test circuit is very similar to the one used to measure open-loop output impedance. Inductor L1 creates closed-loop feedback at DC while allowing for open-loop AC analysis, and capacitor C1 shorts the inverting input to signal source Vin at AC in order to receive the appropriate AC stimulus. As explained by Bruce Trump in his classic blog post, “ Offset Voltage and Open-Loop Gain – they’re cousins ,” you can think of Aol as an offset voltage that changes with DC output voltage. Therefore, to measure Aol, run an AC transfer function over the desired frequency range and plot the magnitude and phase of Vo/Vos. Make sure to match the specified data sheet conditions for power-supply voltage, input common-mode voltage, load resistance and load capacitance. Let’s use this method to test the Aol of the OPA189 . Figure 5: OPA189 Aol result In this case, the op amp’s Aol is modeled very closely to the data sheet spec. The spike in the data sheet’s Aol around 200kHz is caused by the chopping network at the input of the amplifier and is not modeled, although its effect on the nearby magnitude and phase response is. Thanks for reading this fifth installment of the “Trust, but verify” blog series! In the sixth and final installment, I’ll discuss how to measure op amp voltage and current noise. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Learn more about Vos in Bruce Trump’s “ SPICEing Offset Voltage ” blog post. Watch these TI Precision Labs – Op Amps videos: “ Vos and Ib .” “ Bandwidth 1 .”opampportalPrecision Op Amps (Vos<1mV)SPICE blogpampportalOPA189Blog Post: “Trust, but verify” SPICE model accuracy, part 4: open-loop output impedance and small-signal overshoothttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/10/24/trust-but-verify-spice-model-accuracy-part-4-open-loop-output-impedance-and-small-signal-overshootTue, 24 Oct 2017 14:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:04ad4bb7-9423-49ab-ac7b-c1e71cfeef94Ian WilliamsPrevious installments of this blog post series discussed the need for verifying SPICE model accuracy and showed how to measure common-mode rejection ratio (CMRR), offset voltage versus common-mode voltage (Vos vs. Vcm) and slew rate (SR). In part four, I’ll continue putting operational amplifier (op amp) SPICE models to the test by checking their usefulness for small-signal stability analysis. Whether instability rears its ugly head as overshoot and ringing, continuous oscillation, or other more bizarre behavior, it can prove to be a real beast to debug. Thankfully, an accurate SPICE model is a valuable asset in the struggle to solve op amp stability issues. A good model, combined with the powerful analysis tools available in simulation, can help predict and stabilize op amp circuits before they get a chance to cause trouble in the real world. While many different stability compensation methods exist, a thorough discussion of stability compensation is beyond the scope of this post. Instead, I’ll focus on how to verify that an op amp SPICE model is accurate for use in stability analysis by comparing model performance versus the data sheet. If you wish to dive deeper into op amp stability theory and compensation techniques, start by watching our TI Precision Labs – Op Amps video series on stability . Open-loop output impedance The most critical specification to check for accuracy before performing stability analysis is the op amp’s open-loop output impedance, or Zo. At a basic level, you can think of Zo as a complex impedance in the op amp’s small-signal path, which occurs between the open-loop gain stage (Aol) and the output pin. This impedance interacts with the op amp’s Aol, as well as any load and feedback present, to create the circuit’s overall AC response. Figure 1 is a simplified schematic-level view of Zo in an open-loop op amp circuit. Figure 1: Simplified op amp small-signal model If op amp manufacturers do not model Zo accurately, then the overall small-signal AC behavior of an op amp model is incorrect and can’t be used for stability analysis. Thankfully, it’s easy to verify that a model’s Zo matches the data sheet. Figure 2 shows the recommended test circuit. Figure 2: Open-loop output impedance test circuit In this test circuit, inductor L1 creates closed-loop feedback at DC while allowing for open-loop AC analysis, and capacitor C1 shorts the inverting input to ground at AC to prevent the node from floating. AC current source I_TEST back-drives the op amp output, and by measuring the resulting voltage at the output pin, you can determine the output impedance using Ohm’s law. To plot Zo, run an AC transfer function over the desired frequency range and plot the voltage at Vout. Note that many simulators default to showing the results in decibels. If you plot the measurement on a logarithmic scale, Vout is equivalent to ohms. Let’s now test the Zo of the OPA202 , a new precision bipolar amplifier from TI. Figure 3: OPA202 Zo results In this case, the op amp’s Zo is modeled very closely to the data-sheet spec. The output impedance is also very flat (that is, resistive) up to around 1MHz, typical of classic bipolar amplifier designs. Confident that the model’s Zo is correct, let’s now check the rest of the small-signal response. Small-signal overshoot One of the simplest ways to check for op amp stability (both with simulation and in the real world) is to measure the percent overshoot at the output in response to a step or square-wave input. Assuming that the op amp circuit is a second-order system, overshoot can be related to phase margin (and therefore stability) based on their mathematical relationship to each other through the damping factor. Figure 4, taken from the “ Analog Engineer’s Pocket Reference ,” shows this relationship as overshoot increases from zero to 100%. Figure 4: Phase margin vs. percent overshoot You can test small-signal overshoot in both inverting and noninverting configurations, but today I’ll be demonstrating the inverting configuration shown in Figure 5. RF and RI are set to the op amp’s typical load resistance of 2kΩ and configure the closed-loop gain to -1V/V. CF provides compensation of the op amp input capacitance and is set equal to C_CM + C_DIFF, while capacitive load CL is set to 10nF. Vin generates a 5mVpk square wave at 10kHz, ensuring that the op amp shows only small-signal behavior. Figure 5: Small-signal step response test circuit, gain = -1V/V Let’s use this circuit to measure the small-signal overshoot of the OPA202 . To do this, first run a transient analysis over one period, or 100µs, and plot the voltage at Vin and –(Vout). Since this is an inverting amplifier setup, I recommend inverting the output waveform for easier comparison against the input. Figure 6: OPA202 small-signal overshoot, gain = -1V/V, CL = 10nF Equations 1 and 2 calculate the percent overshoot: % overshoot = 100 * [(Vmax – Vfinal) / Vstep] (1) % overshoot = 100 * [(7.11 mV – 5 mV) / 10 mV] = 21.1 % (2) where Vmax is the maximum output voltage, Vfinal is the final settled output voltage and Vstep is the total output step size. Referring back to the chart in Figure 4, a percent overshoot of 21.1% corresponds to roughly 47 degrees of phase margin. One general recommendation for stability is that a circuit should have at least 45 degrees of phase margin, so this just meets that requirement. It’s quite remarkable that the OPA202 is still stable even with a 10nF load! You can repeat this test with different capacitive loads to see how well the OPA202 model matches the data-sheet capacitive load drive spec. Figure 7 gives those results. Figure 7: OPA202 overshoot vs. capacitive load comparison Sweeping CL from approximately 30pF to 25nF, the SPICE model overshoot aligns quite closely with the data-sheet curve, especially at heavier loads. This indicates that the small-signal characteristics of the SPICE model very closely match the real device, and any stability compensation done with simulation will translate well to the real world. Thanks for reading this fourth installment of the “Trust, but verify” blog series! In the next installment, I’ll discuss how to measure open loop gain (A OL ) and input offset voltage (V OS ). If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Learn more about model simulation in Bruce Trump’s “ SPICEing Op Amp Stability ” blog post. Download the Solving Op Amp Stability Issues presentations. Learn more in the OPA202 datasheet . Read these Analog Applications Journal articles: “ Modeling the output impedance of an op amp for stability analysis .” “ Op amp stability and input capacitance .” SaveopampportalOPA202Precision Op Amps (Vos<1mV)SPICE blogpampportalBlog Post: “Trust, but verify” SPICE model accuracy, part 3: slew rate and input clamping diodeshttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/18/trust-but-verify-spice-model-accuracy-part-3-slew-rate-and-input-clamping-diodesMon, 18 Sep 2017 17:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:01fde1b8-0655-4eca-9706-071290f0838dIan WilliamsPrevious installments of this blog post series discussed the need to verify SPICE model accuracy and how to measure common-mode rejection ratio (CMRR) and offset voltage versus common-mode voltage (V OS vs. V CM ). In part 3, I’ll continue by explaining how to verify an operational amplifier (op amp) model’s slew rate, which is a large-signal output response. Slew rate Slew rate is defined as the maximum rate of change of an op amp’s output voltage and is typically given in volts per microsecond (V/µs). Slew rate is a type of output distortion, or nonlinearity. An amplifier in this condition is not behaving linearly where the output voltage equals the input voltage multiplied by the closed-loop gain. Instead, the op amp output voltage changes with a constant slope. This continues until the op amp corrects the difference at its input pins and the amplifier returns to a linear or small-signal operating state. For a more detailed look at slew rate, watch our TI Precision Labs – Op Amps video series on slew rate . One of the most common ways to force an op amp into slew rate limit is to apply a large-signal input step of 100mV or greater, but slew limiting can also occur when trying to output high-amplitude signals at high frequencies. In audio applications, for example, slew limiting distorts sine waves into triangle waves, causing visible (and audible) distortion, as shown in Figure 1. Figure 1: Triangular distortion caused by slew rate limit When using SPICE simulation for audio applications or circuits where large-signal input steps are common (such as those with multiplexers or switches), I recommend verifying the slew rate behavior of your op amp model. Figure 2 shows the recommended test circuit. Figure 2: Slew rate test circuit This circuit places the op amp in a unity-gain buffer configuration and applies a large-signal step to the noninverting input pin. The amplitude of the step should match the test conditions given in your specific op amp’s data sheet. Let’s test the slew rate of the OPA196 , a new e-Trim TM precision amplifier from TI, whose data sheet specifies a 10V input step test condition (Figure 3). To determine the slew rate, measure V OUT and calculate its rate of change as it transitions from 10% to 90% of the total output step. Figure 3: Slew rate test results for the OPA196 Equations 1 through 3 calculate the rising slew rate for the OPA196 : where ΔV is the change in voltage and Δt is the change in time from 10% to 90% of the output transition. In this case, the simulated rising slew rate perfectly matches the data sheet spec of 7.5V/µs! You can repeat this test with a negative input step to measure the falling slew rate. Testing amplifiers with input clamping diodes Testing the slew rate of certain types of amplifiers requires a small tweak to the slew rate test circuit. On most bipolar, high-voltage complementary metal-oxide semiconductor (CMOS) and chopper amplifiers, clamping diodes are present across the op amp input pins. If you try to apply a large-signal step to these devices, the large differential input voltage will cause these diodes to conduct current directly from the noninverting input to the inverting input and output. The result is an incorrect slew rate measurement that’s faster than what the actual device can generate. Figure 4 shows this effect with the OPA1678 , a high-voltage CMOS audio amplifier from TI. Figure 4: Slew rate test results (no input current limit) for the OPA1678 This effect was not evident on the OPA196 , even though it’s a high-voltage CMOS amplifier. That’s because it’s part of TI’s OPA19x family of amplifiers with multiplexer-friendly inputs. The design of the OPA19x family eliminates the need for input clamping diodes, and the amplifiers can handle large differential input voltages without them. Junction field-effect transistor (JFET)-input amplifiers such as the OPA145 also do not exhibit this issue. To test the true slew rate of amplifiers with input clamping diodes, place a current-limiting resistor either in series with the input source or between the inverting input and the output pin. A resistance of 10kΩ does the trick for the vast majority of op amps. Figures 5 and 6 show the modified circuit and test results. Figure 5: Slew rate test circuit with input current limit Figure 6: Slew rate test results with input current limit for the OPA1678 Using Equations 1-3 once more, the rising slew rate of the OPA1678 model is 8.9V/µs – very close to the data sheet spec of 9V/µs. As an alternative to using a large input current-limiting resistor, you can test the slew rate with the amplifier in an inverting configuration. In this case, the input and feedback resistors limit the input current through the input diodes and enable an accurate measurement. Thanks for reading the third installment of the “Trust, but verify” blog series! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Read the Bruce Trump article, “ Slew Rate – the op amp speed limit .” To learn more about op amp input clamping diodes, watch the video, “ TI Precision Labs – Op Amps: Comparator Applications 4 .” Download the TI reference guide, “ TI Precision Designs: Reference Design Single Op Amp Slew Rate Limiter .” SaveopampportalOPA1678Precision Op Amps (Vos<1mV)SPICE blogaudioopampportalOPA145pampportalaudioportalPrecision Op AmpsOPA196Blog Post: Speed up basic circuit design with the analog engineer’s calculatorhttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/07/speed-up-basic-circuit-design-with-the-analog-engineer-s-calculatorThu, 07 Sep 2017 15:30:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:4f7e1a6c-6df6-44a0-901e-bc68de7a9689Jim CattQuick quiz: Can you find the standard 1% resistor values for a voltage divider that comes closest to a divider ratio of V OUT /V IN = .3278, with less than 0.01% error? The answer is 324Ω and 158Ω, with 0.00025% error. Setting up the equation to solve for one of the values in terms of the other is easy. But iterating through multiple standard resistor values is tedious and time consuming, even when using a spreadsheet. The analog engineer’s calculator simplifies this task. This newly developed tool is a companion to the “ Analog Engineer’s Pocket Reference .” Many of you are familiar with this e-book, which covers many fundamental topics in circuit design: unit conversion, components, circuit equations, op amps, printed circuit board (PCB) design, sensors and analog-to-digital converters (ADCs). For those of you who hate memorizing even basic formulas and equations (or more likely, have gotten a little rusty), the pocket reference is an easily accessible source that can save tons of time (unless, of course, you have meticulously indexed your college textbooks). This beta tool contains a collection of simple-to-use calculators that support much of the content in the pocket reference. While it doesn’t address every topic, it does cover the more interesting and complex topics, and constitutes one-stop shopping for many of the simple calculations that you might perform regularly. Figure 1 lists the possible calculations. Figure 1 : Analog engineer’s calculator menu The calculator is especially useful when designing sensor signal-conditioning and data-acquisition systems to monitor voltage, current and temperature. The built-in calculators for amplifiers, data converters and temperature sensors make the task easier and faster. Need to design an input drive circuit for a successive approximation register analog-to-digital converter (SAR ADC)? Use the ADC SAR drive calculator to design the circuit. As Figure 2 shows, simply select the input type (single ended, differential, etc.); enter the ADC resolution, sampling cap value, full-scale input range and acquisition time; and click OK to see the associated resistor-capacitor circuit values, as well as other parameters. Figure 2 : ADC SAR drive calculator Analog designers often need to make cascaded noise calculations when selecting circuit components to meet target specifications. Setting up signal-chain noise calculations can be tedious, but the calculator enables quick computations using only a few input parameters, as shown in Figure 3. Figure 3 : ADC plus signal-chain noise calculation How many times have you designed a simple inverting or noninverting gain stage and wanted to get as close as possible to your target gain using only standard 1% resistors? You probably know that breaking up the feedback resistance into two or more discrete values reduces the error caused by the resistor tolerances. Selecting the 1% resistor values and calculating the actual gain and gain error isn’t difficult, just tedious. You have more important things to do. Use the amplifier gain resistor calculator to speed up the task (Figure 4). Figure 4 : Gain resistor calculator Perhaps you need to calculate the inductance and capacitance of a section of PCB trace. Use the microstrip calculator to quickly find these values by entering a few trace parameters (Figure 5). Figure 5 : Microstrip calculator These examples represent just a few of the often-used analog design calculations, aggregated into a single tool that you can place on your desktop and access offline. No more bouncing between bookmarks on the web. Download the “ Analog Engineer’s Pocket Reference ” and test out the analog engineer’s calculator to begin exploring the useful features that will speed up basic design tasks and save you valuable time. Sign in and comment below if you have any feedback or suggestions about the calculator. Additional resources Learn more about TI’s amplifier and data converter portfolios.opampportalADC blogspadcportalprecision op amp blogspampportaldcportalSAR ADC blogsBlog Post: “Trust, but verify” SPICE model accuracy, part 2: input offset voltage vs. input common-mode voltagehttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/05/trust-but-verify-spice-model-accuracy-part-2-input-offset-voltage-vs-input-common-mode-voltageTue, 05 Sep 2017 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:8bd27877-ca3b-47f5-924f-eda542545bc1Ian WilliamsIt’s no secret that low-voltage rail-to-rail input operational amplifiers (op amps) are gradually taking the place of traditional high-voltage amplifiers in many precision applications. Rail-to-rail input amplifiers are extremely useful, since their linear input-voltage range spans the entire power-supply voltage range (or even beyond). They traditionally achieve this span through the use of two pairs of input transistors instead of one pair, but you should be mindful of new design challenges that this topology creates. One challenge is the change in the op amp’s input offset voltage (V OS ) when the amplifier input stage crosses over from one pair of transistors to another. This phenomenon is often called input crossover distortion. V OS is an important performance characteristic of a precision op amp, and many systems must calibrate out the initial offset voltage to meet their performance goals. Any changes to V OS , whether caused by changes in input common-mode voltage (V CM ), temperature or other variables, are highly undesirable and can throw off a system’s total error performance. Figure 1 gives an example of V OS changing dramatically with increased V CM . Figure 1: V OS vs. V CM When using SPICE simulation for rail-to-rail input amplifier designs, it’s wise to check that the V OS vs. V CM behavior of your models matches the real devices. Figure 2 shows the recommended test circuit. Figure 2: V OS vs. V CM test circuit This simple circuit places the op amp in a unity-gain buffer configuration to prevent output swing limitation issues, then sweeps V CM to determine the change in V OS . To plot V OS vs. V CM , run a DC transfer characteristic while stepping V CM across the entire supply voltage range and measure V OS across the op amp input pins as shown in Figure 2. Let’s use this method to test the response of the OPA388 , a new zero-crossover precision amplifier from TI that uses a charge pump in its input stage to achieve true rail-to-rail performance using only a single transistor pair. This eliminates the input crossover distortion found in traditional rail-to-rail input op amps. See Figure 3. Figure 3: V OS vs. V CM results of the OPA388 The simulated results match the responses of the three test devices given in the OPA388 data sheet very closely, with a change of less than 1μV over the entire V CM range. Let’s use the same test circuit to check the response of the OPA2325 , another zero-crossover precision amplifier from TI. See Figure 4. Figure 4: V OS vs. V CM results for the OPA2325 Again, the simulated results match the real silicon very well. Keep in mind that while the simulation model looks like it has higher offset than the real silicon, all of the test devices measured in this plot had a V OS lower than the typical spec of 40μV, while the SPICE model was designed to match the typical. Thanks for reading the second installment of the “Trust, but verify” blog series ! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Download the TI tech note, “ Zero-Crossover Amplifiers: Features and Benefits .” Watch these videos to learn more about V OS vs. V CM : “ TI Precision Labs – Op Amps: Low Distortion Design 2 .” “ TI Precision Labs – Op Amps: Input and Output Limitations 1 .”OPA2325opampportalPrecision Op Amps (Vos<1mV)OPA388SPICE blogpampportalPrecision Op AmpsUltra-Low-Power Op Amps (IQ<=10µA)Blog Post: Transimpedance amplifier designs for high-performance, cost-sensitive smoke detector applicationshttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/08/10/transimpedance-amplifier-designs-for-high-performance-cost-sensitive-smoke-detector-applicationsThu, 10 Aug 2017 14:37:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:58ef0f76-6fa0-4b21-9ed2-4f19253d6531Amanda WeiseThis post is co-authored by Collin Wells . Photodiode-based light sensing is a common technique where operational amplifiers (op amps) are used to condition a photodiode sensor for a wide range of applications. An example of this is in smoke detectors, where a photoelectric smoke alarm is used to identify the presence of smoke in the chamber. A photodiode sensor produces a current proportional to the light level presented to it. Depending on the application, the photodiode is operated in either a photovoltaic or photoconductive mode; each has its own merits, which Bruce Trump discussed in detail in this post from his blog, The Signal . In a smoke detector system, the photodiode operates in a photoconductive mode, meaning you will typically use a transimpedance amplifier to amplify the photodiode current. In photoconductive mode, the photodiode is held at a zero-volt (Figure 1a) or reverse voltage bias (Figure 1b), preventing it from forward biasing. Figure 1: Photodiode in photoconductive mode with zero-volt bias (a); or reverse-voltage bias (b) Equation 1 calculates the direct current (DC) transfer function for the circuits shown in Figure 1 (note that the photodiode current (i D ) is flowing away from the op-amp inverting node): The three-step process outlined in John Caldwell’s series on transimpedance amplifiers (see part 3, “ What op amp bandwidth do I need ?”) determines the minimum required op-amp gain bandwidth for transimpedance configurations. The minimum bandwidth is based on the required transimpedance gain and signal bandwidth, along with the total capacitance presented to the inverting node of the op amp. The diode capacitance often dominates the inverting-node capacitance, but don’t forget to include the effects of the op-amp input capacitance. We summarized the three steps explained in John’s posts here for quick reference. 1. Choose the maximum feedback capacitance (C F ) based on the feedback resistor (R F ) and the signal -3dB bandwidth (f P ) (Equation 2): 2. Calculate the total capacitance (C IN ) at the inverting input of the amplifier. For the circuits shown in Figure 1, this is equal to Equation 3: where C J is the diode junction capacitance, C D is the op-amp differential input capacitance and C CM2 is the op-amp inverting input common-mode input capacitance. 3. Calculate the minimum required op-amp gain bandwidth product (f GBW ) (Equation 4): By following these three simple steps, you can avoid many of the stability and performance issues commonly associated with transimpedance amplifier circuits by selecting an amplifier with sufficient bandwidth to perform the required transimpedance gain at the desired signal bandwidth. Op amps with GBWs between 1 - 20MHz are well suited for smoke detector applications because they are able to amplify the low-level signals in the system to sufficient levels while maintaining stability. Along with meeting bandwidth requirements, the op amp must also meet the system’s DC accuracy requirements. The most important DC specification in many transimpedance applications is the input bias current (i B ) of the op amp. i B will directly sum or subtract with the input signal current, which can cause large errors depending on the magnitude of i B compared to the signal current. In smoke detector applications, this DC voltage enables system designers to set the thresholds for the amount of smoke detected before an action is taken in the system. The example shown in Figure 2 uses a 5MΩ resistor to apply a 5MV/A gain to a 100nA full-scale input current. With the input bias current set to 0A, the full-scale output voltage is 500mV – which is expected based on the transfer function in Equation 1. The circuit on the right in Figure 2 displays the effects of the same circuit with an op amp i B of 10nA. In this case, the output voltage is 450mV, which shows that the 10nA input bias current caused a 50mV (or 10%) error from the ideal 500mV output signal. Figure 2: Input bias current effects in transimpedance amplifier circuits Equation 5 calculates the percentage error of the full-scale range (%FSR) based on the full-scale input current (i IN_FS ) and the op amp’s i B : The TLV6001 device is part of a family of high-performance general-purpose amplifiers for a wide variety of cost-conscious transimpedance applications, such a smoke detectors. This is due to the op amps balance between GBW (1 MHz), low Iq (100 µA), low input bias current (1 pA) and low input capacitance (6pF). Other key features that make this family attractive for system designers in smoke detectors are the RRIO swings and the EMI hardened inputs that help reject interference from extrinsic noise sources. Table 1 lists different transimpedance gain and bandwidth combinations for the TLV6001 based on Equations 1 through 4. Be sure to keep the total input capacitance below the maximum input capacitance (C IN­_MAX ) to avoid stability issues. Table 1 : Quick design calculator for TLV6001 transimpedance applications Figure 3 shows the simulated step-response results for a 1MV/A gain and 50kHz bandwidth with the maximum 54pF of input capacitance from the photodiode. The output overshoot and ringing are minimal, indicating a stable design. Figure 3: TLV6001 step-response results; gain = 1MV/A, bandwidth = 50kHz Smoke detector applications use op amps in the transimpedance configuration to amplify low-level photodiode currents. Designing the transimpedance circuit for smoke detectors can be simplified to a few easy steps. First, follow the three steps from John’s blog posts to select the required op-amp bandwidth. Then sort the remaining results to find a device with an i B specification that meets the system’s DC requirements. The TLV600x devices highlighted in this blog are a great family of products to design with for cost-sensitive transimpedance applications. Do you have questions about op amp designs in smoke detectors? Log in and leave a comment below letting us know your experience with transimpedance configurations or any questions you have. Additional resources Find commonly used analog design formulas in the “ Analog Engineer’s Pocket Reference ” e-book. Download The Signal e-book for 32 op amp design lessons that will make you a better engineer. See TI’s portfolio of performance op amps for cost-conscious applications. Read the other two parts in John Caldwell’s three-part blog series on transimpedance circuits: “ Part I – What op amp bandwidth do I need? (Transimpedance Amplifiers) .” “ Part II – What op amp bandwidth do I need? (Transimpedance Amplifiers) .”opampportalTransimpedance Amplifier Circuitgeneral purpose op amp blogscircuit0020Precision Op Amps (Vos<1mV)TLV6001General Purpose Amplifiergeneral-purpose op ampspampportalPrecision Op AmpsSmoke and Heat DetectorBlog Post: “Trust, but verify” SPICE model accuracy, part 1: common-mode rejection ratiohttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/07/27/trust-but-verify-spice-model-accuracy-part-1-common-mode-rejection-ratio-cmrrThu, 27 Jul 2017 15:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:384ba23f-a49f-4a4a-84b5-9fe7bfc65aceIan WilliamsSPICE simulation is an immensely valuable tool that allows engineers to have high confidence in their analog designs before ever stepping foot in a lab. While many of us live and die by our simulation results, have you ever stopped to question if your favorite operational amplifier’s (op amp) SPICE model matches the specs promised by its data sheet? Our competitors’ models can often be inaccurate or oversimplified, as shown in the closed-loop output impedance (Zout) comparison in Figure 1. In this case, it’s easy to see the difference between what’s promised and what’s measured. This difference can have a big impact on your designs, potentially causing some unwanted surprises when you fire up your circuit in the lab for the first time. Figure 1: Competitor SPICE model Zout comparison TI is proud to have some of the most accurate SPICE models in the industry. That being said, it’s still a good practice to “trust, but verify” that the models used in your circuit are up to the task. In this blog series , I will show you the recommended test circuits for many of the most critical op amp parameters, starting with common-mode rejection ratio (CMRR). An op amp’s CMRR is formally defined as the ratio of its common-mode gain to its differential-mode gain. In practical terms, the CMRR spec tells you how much additional offset voltage is generated at the op amp’s input when the input common-mode voltage changes. A high CMRR is desirable, meaning that you’ll see less additional offset voltage. CMRR also changes over frequency, so TI data sheets provide a curve representing typical CMRR over frequency, as shown in Figure 2. For a deeper discussion of CMRR and its impact on circuit performance, watch our TI Precision Labs – Op Amps video on CMRR . Figure 2: OPA2187 CMRR vs. frequency I’ll provide two different approaches for measuring op amp CMRR in simulation. Both of these methods were developed by my colleague Zak Kaye , who is well on his way to becoming an analog guru! Modified differential amplifier method The first method is a modification of the classic differential amplifier circuit, shown in Figure 3, that will test the CMRR of the OPA2187 , a new chopper amplifier from TI with 140dB of common-mode rejection. Figure 3: CMRR test circuit (modified differential amplifier) This circuit takes advantage of a SPICE simulator’s ability to isolate nodes using voltage-controlled voltage sources. We’ve strategically placed these sources at the op amp input and output to isolate the feedback network from the op amp’s input and output impedance, allowing for better extraction of the CMRR response. To plot the CMRR, simply run an AC transfer function and use post-processing to create a curve for (Vcm / Vos). Figure 4 shows the results for this configuration compared to the OPA2187 data sheet. Figure 4: Modified differential amplifier results The circuit extracted CMRR fairly well, but the simulation doesn’t match the data sheet that closely at high frequencies. Is that because the model is inaccurate, or are there limitations with the test circuit? Let’s try another circuit configuration and find out. Open-loop method The second test method, shown in Figure 5, uses an open-loop approach. Impossibly large inductors create closed-loop feedback at DC while still allowing for open-loop AC analysis, with two instances of the op amp separately measuring the common- and differential-mode gain. Figure 5: CMRR test circuit (open-loop method) To plot CMRR, run an AC transfer function and use post-processing to create a curve for (ADM / ACM). Figure 6 gives the results for this configuration compared to the OPA2187 data sheet. Figure 6: Open-loop method results As you can see, the open-loop results match the data sheet very closely due to the true isolation of the common- and differential-mode gains of the op amp. I recommend this approach whenever possible. INA1650 CMRR measurement Finally, I’d like to show an easy way to measure CMRR on the INA1650 , our new high common-mode rejection audio line receiver. Because of the highly integrated design of the INA1650 , you can simply connect each input to the same AC source, run an AC transfer function, and invert the output in order to extract CMRR, as shown in Figures 7 and 8. Figure 7: INA1650 CMRR test circuit Figure 8: INA1650 CMRR test results Thanks for reading this first installment of the “trust, but verify” blog series! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the E2E™ Simulation Models forum . Additional resources Download these TI TechNotes to learn more about TI’s op amps: “ Zero-Drift Amplifiers: Features and Benefits .” “ Offset Correction Methods: Laser Trim, e-Trim™ and Chopper .”opampportalPrecision Op Amps (Vos<1mV)INA1650SPICE blogOPA2187pampportalPrecision Op AmpsBlog Post: Zero out your system error with zero drift, zero crossover and zero hasslehttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/04/21/zero-out-your-system-error-with-zero-drift-zero-crossover-and-zero-hassleFri, 21 Apr 2017 15:30:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:ee0d4a8c-2fa8-49fa-bf3e-3439e31f4571TamaraThis post is co-authored by Richard Barthel and Errol Leon . In applications such as position sensors, data-acquisition systems and resistance temperature detectors (RTDs), it is important to design with high precision in mind. In many cases, designing with precision integrated circuits (ICs) reduces signal-chain complexity, lowers the external component count, and minimizes board space and bill of materials (BOM) costs. The inaccuracies of one device may propagate through with the inaccuracies of another device, resulting in undesirable and unpredictable errors. In the case of a buffer-configured operational amplifier (op amp) at the output of a digital-to-analog converter (DAC), it’s crucial that your DAC and your op amp are precision devices for an accurate output. A traditional rail-to-rail complementary metal-oxide semiconductor (CMOS) amplifier architecture includes two differential pairs, PMOS (blue) and NMOS (red), shown in Figure 1. Together, these two transistor pairs span the entire input common-mode voltage range. When one transistor pair takes over from the other, however, a unique and nonlinear phenomenon known as “input crossover distortion” occurs due to the intrinsic input offset voltage of each of the two input differential pairs, shown in Figure 2. Figure 1: Traditional rail-to-rail CMOS amplifier architecture Figure 2: Input offset voltage vs. common-mode voltage When you connect a traditional rail-to-rail CMOS op amp at the output of a high-precision DAC, the crossover distortion will introduce an error and result in a drastic increase in integral nonlinearity (INL). This may cause the signal to deviate several least significant bits (LSBs) from its ideal value. Now, what does 1LSB mean? Equation 1 is a simple equation to calculate LSB: where N is the DAC’s number of bits. The DAC8830 is a 16-bit DAC. If the voltage reference is V ref = 5V, then: So to deviate more than 1LSB means that you can have more than 76.3µV of error at your output. This can be detrimental to many precision applications, like critical systems where failure has the potential to negatively impact customers’ end products. So how do you fix this? Enter zero crossover! You can span the entire input common-mode voltage range by using a zero-crossover op amp such as the OPA388 . The zero-crossover topology uses an internal regulated voltage charge pump to increase the positive supply voltage and thus achieve linear operation with input common-mode voltages all the way to its rails with a single input transistor pair, shown in Figure 3. This results in true rail-to-rail input operation without a crossover region, and thus no crossover distortion. If you were to connect this kind of op amp at the output of a DAC, the op amp does not introduce an error within the common-mode region (1V to 2V below the positive rail) like a traditional rail-to-rail CMOS device. Figure 3: Zero-crossover amplifier architecture In Figure 4, the black curve describes the output of a traditional rail-to-rail CMOS op amp ( OPA340 ) at the output of a DAC ( DAC8830 ), while the red curve describes the output of a zero-crossover op amp ( OPA388 ) with the same DAC8830 . As you can see, the output of the DAC8830 + OPA388 does not suffer from the distortion that is easily visible in the DAC8830 + OPA340 output curve. The High-Precision Reference Design for Buffering a DAC Signal describes this output in greater detail. Figure 4: INL comparison of rail-to-rail CMOS OPA340 and zero-crossover OPA388 Let’s put this reference design into perspective and use it in an application such as an MRI machine. An MRI uses a powerful magnetic field to produce detailed 2-D and 3-D pictures of the human body to diagnose and/or monitor several health conditions. Unacceptably distorted signals that exceed the error budget in any way can potentially impair the quality of the images. The OPA388 is the industry’s first op amp to employ zero-crossover and zero-drift technology. Zero-drift op amps have an internal self-correcting circuit that produces ultra-low input offset voltage (V OS ) and near-zero input offset voltage drift over time and temperature (dV OS /dT). The technology also delivers other advantages, including no 1/f noise (flicker noise), low broadband noise (white noise) and low output distortion, which can help increase system reliability in harsh environments. Take a swimming pool for instance – pH pool testers and monitoring systems must withstand changes in the environment’s temperature to correctly determine the deficit or excess of chlorine. Since most pools are placed outside, the environment’s temperature can vary many degrees between a cold winter’s day and a hot summer’s day. Offset voltage will change with temperature deviations, introducing error, so it is crucial to select an op amp with low offset voltage drift to support system reliability through these changes. To assure high performance, high precision and high accuracy, carefully select parts for your design. Make sure that you understand your system and what you can afford in terms of error, and only then sift through TI’s diverse portfolio for your ultimate solution. Additional resources Download the zero-drift and zero-crossover Tech Notes. Visit the OPA388 and DAC8830 datasheets to learn more.opampportalprecision op amp blogsOPA388DAC8830pampportalPrecision Op Ampsopa340Blog Post: Threading the needle: new op amp delivers exceptional precision, accuracyhttp://e2e.ti.com/blogs_/b/thinkinnovate/archive/2017/03/06/threading-the-needle-new-op-amp-delivers-exceptional-precision-accuracyMon, 06 Mar 2017 14:29:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:36d7459d-25b8-4463-b849-a65078f9f43fThink. Innovate. contributorPrecision affects every part of our lives – from ordering lunch meat at the deli to calibrating delicate lab equipment. And a new high-precision operational amplifier – a circuit that magnifies tiny electric currents so that they can be measured and managed – offers an exceptional level of accuracy in industrial, automotive, medical, personal electronics, and test-and-measurement equipment. “The performance on this device is so precise that it’s like measuring a dime on the Empire State Building,” said Richard Barthel, a systems engineer who worked on the team that developed the OPA388 . “It takes something very small and gets it right on target.” Operational amplifiers – or op amps – are links between sensors that measure analog signals such as pressure, temperature and flow and the digital brains behind technologies that are so integral to our everyday lives. Sensors pick up analog signals from the environment around us that in many cases are measured in the millionths of a volt – too small to be useful for the circuits that convert them to digital signals. The job of op amps is to boost those signals to higher voltages that can then be measured, interpreted and managed by computers. In that amplification process, any variation in the signal gets progressively more distorted as it works its way through the subsequent signal chain. That, in turn, affects the precision of the final measurements produced by a piece of equipment, said Ying Zhou, product marketing manager for the device. Our zero-drift and zero-crossover technologies – which our newest op amp combines in one device for the first time – correct for any noise and errors in the signals and remove the need for designers to add discrete calibration circuits to the systems they create. Combining these technologies will lead to improvements in the accuracy of measurements in applications ranging from electronic scales to heart-rate monitors and pressure sensors. For example, the device’s precision is beneficial for equipment such as: Gauges used in CT scan machines . This medical equipment requires a smooth and consistent movement so the weight and weight distribution of patients can be measured precisely, a critical factor in accurate diagnosis and treatment. Construction equipment . Contractors, civil engineers or other workers will be able to measure elevation and distance with pinpoint precision during building construction, which could increase structural integrity. Weigh scales. Achieve more precise weight measurements – whether you’re ordering a few slices of lunch meat at a deli or checking the weight of a fully loaded semi-truck. Medical lab equipment. Smaller equipment and more precise diagnoses could mean less time in the clinic and more accurate diagnoses for patients. “Precision affects everybody,” Richard said. “The OPA388 takes very small measurements and gets them right on target with high accuracy and resolution. It threads the needle.” If you are interested in learning more, read these TI Tech Notes regarding the features and benefits of zero-drift and zero-crossover technologies.opampportalPrecision Op Amps (Vos<1mV)precision op amp blogsOPA388test measurementpampportalPrecision Op AmpsBlog Post: Designing a discrete wide-bandwidth, cost-sensitive instrumentation amplifierhttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/02/03/designing-a-discrete-wide-bandwidth-cost-sensitive-instrumentation-amplifierFri, 03 Feb 2017 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:b40acc93-226b-4961-a180-706b30889dfdCole MaciasIn this post, I’ll show how to design a cost-optimized, discrete, wide-bandwidth instrumentation amplifier using the TLV3544 . Instrumentation amplifiers are used for their high input impedance and ability to convert differential voltages to single-ended voltages. Fast current sensing, precision data acquisition, vibration analysis, microphone pre-amplification, ADC drivers and medical instrumentation are all applications that need instrumentation amplifiers with wide bandwidth. TLV354x devices provide a rail-to-rail input/output, 200MHz unity gain bandwidth (GBW) and 150V/µs slew rate, which is designed for the applications I just mentioned. Figure 1 shows the standard three- operational amplifier (op amp) topology. Figure 1: Discrete three-op-amp topology using the TLV3544 The input stage uses dual noninverting amplifiers that enable high impedance at both inputs, whose gain is defined by RF1 = RF2 and RG1. The output stage consists of a difference amplifier with a low impedance output, whose gain is set by R2 = R4 and R1 = R3. The reference voltage, input stage gain and output stage gain define the output voltage, shown in Equation 1: Note that the tolerance of the resistors in the instrumentation amplifier will negatively affect the CMRR and gain error of the circuit. That is why there is a cost and performance trade-off between discrete and integrated instrumentation amplifiers. The bandwidth of instrumentation amplifiers is bounded by three characteristics: the open-loop gain (Aol) of the op amp, the noise gain (Gn) and filtering. Both Aol and Gn are covered in the TI Precision Labs training series on bandwidth (see part 3, “ TI Precision Labs – Op Amps: Bandwidth 3 ” – viewing requires a myTI login). The output-stage difference amplifier’s noise gain determines the circuit’s bandwidth. The TLV3544 , like many high-speed amplifiers, will have stability issues if the feedback resistors are too large. Figure 2 simulates the effects of 200Ω feedback resistors versus 500Ω feedback resistors. Note that 45 degrees of phase margin (PM) is necessary for stable operation. Increasing the resistor values allows for lower power consumption and larger RG1 values to set the gain. Figure 2: TINA-TI ™ software frequency response showing AC peaking, PM and bandwidth with 200Ω (54 degrees PM) versus 500Ω (26 degrees PM) feedback resistors Since the instrumentation amplifier is discrete, you have access to the feedback paths, which allows you to compensate for AC peaking. By placing capacitors in the feedback path, you introduce multiple poles that create a low-pass filter and attenuate the peaking. Equation 2 calculates the -3dB frequency (f p ) of this low-pass filter: Figure 3 shows the schematic and frequency response. Remember that adding a filter will affect the overall bandwidth of the circuit. Figure 3: TINA-TI software schematic and frequency response of compensated feedback paths Since this instrumentation amplifier topology only requires three op amps, the fourth op amp provided by the TLV3544 can serve as a reference buffer for single-supply systems or as an integrator for high-pass filtering of the input. Thus, you can use the TLV3544 to make a discrete wide-bandwidth instrumentation amplifier that is ideal for cost-optimized, high-precision, wide-bandwidth applications. Have questions or comments about other design considerations for discrete instrumentation amplifiers? Log in and leave a comment. Additional resources Watch more than 40 on-demand videos about topics such as bandwidth and stability on TI Precision Labs . Read Pete Semig’s blog series on V OUT vs. V CM limitations, “Instrumentation amplifier V CM vs. V OUT plots: part 1 , part 2 , part 3 ,” to avoid common pitfalls when using instrumentation amplifiers. See TI’s portfolio of performance op amps for cost-conscious applications. Find commonly used analog design formulas in our wildly popular and free Analog Engineer’s Pocket Reference e-book . Read more blogs about precision amplifiers . Learn about TI’s entire portfolio of amplifier ICs and explore technical resources.opampportalfitness machinesInstrumentation AmplifiersWireless Patient MonitorTLV3544analog bloggeneral-purpose op ampsWearable Fitness & Activity Monitorpampportaltina-tiTLV blog seriesBlog Post: Achieve big board-size reductions with tiny, precision op ampshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/01/20/achieve-big-board-size-reductions-with-tiny-precision-op-ampsFri, 20 Jan 2017 16:27:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:ff00257d-26ee-4fe6-aace-c17e39dd8be1Ying ZHOUElectronics such as smartphones, tablets, notebooks and wearable products are becoming more multifunctional, smaller and slimmer. Achieving higher functionality in smaller form factors requires extremely tiny ICs. Many times, different package types help reduce size and solve various design challenges. Take operational amplifiers (op amps), for example; wafer chip-scale packages (WCSP) typically enable the smallest possible footprint for optical modules and wearables, while flat no-lead packages such as quad flat no-lead (QFN) or dual flat no-lead (DFN) inspire differentiated audio functionality in personal electronics where small size, high performance, and easy testing and tuning are essential. Small-outline transistor (SOT) packages like SOT553 are suitable for emerging industrial applications such as field transmitters, which require a wide supply but a small footprint in a user-friendly leaded package. The rapid evolution of mobile devices is driving package technology innovations, but the smallest possible package will always be the size of the die itself. The WCSP is a type of package that can be almost as small as the die, so it’s desirable in applications like medical diagnostics, fitness monitoring and handheld electronic devices. The OPA2376 in WCSP (1.11mm by 2.15mm by 0.625mm) is a device commonly used for signal conditioning purpose in the areas mentioned above. Audio in portable devices is another example where tiny packages are inspired by the consumer needs. Traditionally in the professional audio world, audiophiles like to use op amps in the dual inline package (DIP), which is more DIY-friendly for audio designers. The OPA2134 from the legacy Burr-Brown TM audio portfolio, is such an example with great audio reputation. However, with the increased demand of high-quality audio in portable devices, tiny-package high-performance audio ICs are required in order to enable the high-fidelity sound in a space-constraint design. With the implementation of the DFN package on the OPA1652 (Figure 1), a performance upgrade to the OPA2134 , you will find a high-performance current-to-voltage converter with great sound quality for cost-optimized portable audio equipment, smartphones and gaming motherboards. Figure 2 shows the total harmonics distortion and noise (THD+N) performance of the OPA1652 and OPA2134 with common-mode impedance mismatching. Figure 1: OPA1652 (DFN) versus OPA2134 (PDIP) Figure 2: THD+N test in a gain of 100: the OPA1652 vs. the OPA2134 A tiny package may not be the only consideration for wearable devices. These types of applications typically spend most of their time in sleep mode until they are needed to measure biometric data. This requirement has inspired op amps with a shutdown feature in tiny QFN packages. For applications where power consumption is a vital concern, tiny packaging helps enable product innovation, because you can achieve both low power and a small form factor. The OPA2316S in a X2QFN (1.5mm by 2mm by 0.4mm) operates down to 1.8V with a wide bandwidth of 10MHz, as well as rail-to-rail performance, making it suitable for battery-powered designs. Some industrial factory automation applications also require a tiny package. Often, designers prefer devices with small-outline lead packages that are easy to prototype, layout and swap with pin-to-pin-compatible ICs. The OPA171 is one of the first micropower 36V op amps offered in both a single SOT553 (1.6mm x 1.6 mm) package and a dual, very-thin shrink small outline package (VSSOP) (2.0mm x 3.1mm), providing an optimized combination of low cost and performance for applications such as tracking amplifiers in power modules, transducer amplifiers and battery-powered instruments.As a fundamental building block for a signal chain, op amps must keep pace with the emerging electronic design trends. TI has developed products in some of the smallest packages available, including WCSP, QFN/DFN and SOT553. With innovations in design and technologies, more tiny-package, low-power precision op amps are on the way. Browse TI’s tiny op amp portfolio to find the best one for your next project. Additional resources Read the Analog Applications Journal article, “ Distortion and source impedance in JFET-input op amps .” Learn more about the TI Design, “ A High-Fidelity Headphone Amplifier for Current Output Audio DACs Reference Design ” Visit the “ Headphone Amplifier for Voltage-Output Audio DACs Reference Design ”opampportalOPA1652OPA171fitness machinesBlood Glucose Monitorsprecision op amp blogstest measurementwearablespampportalOPA2134Precision Op AmpsOPA2376Blog Post: Multiplexers: not so simplehttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/11/18/multiplexers-not-so-simpleFri, 18 Nov 2016 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:00783dc3-148c-4bb7-8026-d9913171ee82Evan SawyerIt’s simple to design a multiplexer (or mux for short) into a signal chain, right? After all, the device simply funnels multiple signals into a data converter. In reality, a mux can significantly impact the performance of a signal chain in a variety of ways. For example, the on-capacitance can cause crosstalk between channels. Signal- and temperature-dependent variations in the on-resistance can introduce signal distortion. Together, the capacitance and resistance of the mux can limit signal bandwidth. Charge injection can introduce transient errors when the mux switches channels and impact settling time at the output. To optimize signal-chain performance, it is important to understand these examples as well as numerous other ways that a mux can impact a signal, especially because multiplexers are optimized for different performance characteristics, and thus for different applications. Figure 1 is an example circuit containing a mux with its output connected to an inverting operation amplifier ( op amp ). Figure 1: A mux connected to an inverting amplifier introduces gain error This circuit is one of many common mux configurations in a signal chain, but as we will discover, this design will lead to significant signal gain error. Assuming that the op amp is ideal (no offset, bias current, input/output limitations, etc.), Equation 1 expresses the signal gain as: Because the MUX36S08 is not an ideal mux and has internal capacitance as well as an on-resistance of 125Ω, Equation 2 expresses the effective gain of the system as: The calculated signal gain in Equation 2 poses a huge problem if the output of the op amp is connected to a data converter designed to receive the full gain, as nearly 40% of the converter’s range would not be utilized. This equation doesn’t even take into account the on-resistance variation that occurs from changes in temperature, signal voltage or the voltage applied to the supplies. Figure 2 shows one of the on-resistance curves for the MUX36S08 . You can see that the resistance changes based on the temperature as well as the applied signal (the source or drain voltage). The curve that results from changing the signal voltage is known as on-resistance flatness, which can introduce nonlinearity and gain variation. Subjecting the circuit in Figure 1 to a full ±18V sinusoidal signal and temperatures from -40°C to 125°C, the on-resistance of the mux can vary from approximately 75Ω to 250Ω, resulting in an effective gain ranging from -0.44 to -0.73. Figure 2: On-resistance vs. source or drain voltage Luckily, you can effectively ignore the on-resistance of the multiplexer through very simple design precautions. Figure 3 shows the output of the mux connected to an op amp configured as a buffer. The high input impedance of the op amp eliminates any gain error the system would otherwise experience. Figure 3: A mux connected to a buffer effectively eliminates gain error caused by the mux on-resistance As a reminder, the effect of on-resistance on signal gain is only one of the many ways that a multiplexer can impact the performance of a system. When you’re ready to learn about the other ways a mux can add error and distortion to a signal as well as how to mitigate the impact a mux has on signal chain performance, check out the new TI Precision Labs training series covering multiplexers. Additional resources Explore more than 40 hands-on trainings and lab videos from TI Precision Labs . Read a blog about whether a low-leakage multiplexer really matters in a high-impedance PLC system.opampportalMultiplexer/Demultiplexer (Mux/Demux)MUX36S08pampportalBlog Post: How to fix your simulations when the macromodel’s voltage noise doesn’t match the datasheethttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/09/30/how-to-fix-your-simulations-when-the-macromodels-voltage-noise-doesnt-match-the-datasheetFri, 30 Sep 2016 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:217bde76-4214-4ac5-9b85-e93db23bd3e8Cole MaciasWhen responding to questions posted on TI E2E™ Community forums , we frequently run simulations using TINA-TI ™ software, a SPICE-based simulation program. Since we are always in the process of updating our simulation models, we sometimes run across SPICE models that are old, outdated or incorrect when modeling performance parameters. One recent example involves the voltage-noise density of the OPA2333 macromodel. Unfortunately, we found that the model’s voltage-noise density curve was less than that given in the data sheet. So in this blog post, I will show you how to verify an operational amplifier’s (op amp) voltage-noise density curve and correct it if necessary. First, you need to know how to generate a voltage-noise density curve using TINA-TI software. In this example, I will use the OPA2333 macromodel and schematic shown in Figure 1. Figure 1: TINA-TI test bench for voltage-noise density The output noise in this configuration uses the op amp with no gain, filtering or other factors that would change the voltage noise over frequency. To simulate the output noise, select Analysis > Noise Analysis, and tick the Output Noise check box shown in Figure 2. Figure 2: How to find noise analysis for output noise Figure 3: Simulated OPA2333 voltage noise Figure 4: OPA2333 voltage noise according to the data sheet You can add noise to the macromodel by inserting a voltage-noise source in the schematic. To get the voltage-noise source, go to File > Open Examples. Select the Noise Sources folder and open the TINA Noise Sources.TSC file shown in Figure 5. Figure 5: Finding the voltage-noise source and equivalent op amp noise model Now, copy and paste the voltage-noise source from the noise-source schematic into the testing schematic and add it to the noninverting input, as shown in Figure 6. Figure 6: Testing schematic for input-voltage noise with a voltage-noise source Double-click on the noise source and select Enter Macro. A tab will open showing the netlist of the voltage source; see Figure 8. Since the OPA333 is a chopper op amp and has no flicker noise, you do not want to add flicker noise to the voltage source. Find the parameters NLF and FLW and change them to 0 and 0.1, respectively. This sets the flicker noise to 0 nV/ √ Hz at 0.1Hz. You will also need to adjust the broadband noise so that the root sum square (RSS) noise is equal to the value you want. In the netlist, this value is represented by the parameter NVR. In Figure 7, I solve for the desired broadband voltage. Figure 8 shows the updated parameters inside the netlist. Figure 7: Solving for the broadband voltage-noise value Figure 8: Netlist of voltage-noise source Running the noise analysis again (Figure 9), you can see that the voltage noise is now the same value shown in Figure 4. Figure 9: Corrected op amp voltage noise Note that this process only works when your macromodel’s voltage-noise density curve is less than that given in the data sheet because RSS noise can only be added. If your macromodel’s voltage-noise density curve is more than that given in the datasheet, the TI Precision Labs on-demand training series includes a short video on how to create your own accurate macromodel for noise simulations. You can find the video here *. Remember, trust your SPICE models, but verify that they are correct. Simulations can take you far, but they need to simulate correctly to produce meaningful results. Additional resources Learn more about noise , how to calculate it and how to simulate it correctly with our online training series, TI Precision Labs – Op Amps. Learn how to trust and verify your SPICE macromodels in Tim Green’s blog post, “ SPICE op amp macromodels: ‘Trust but verify’ .” Find commonly used analog design formulas in our wildly popular and free “ Analog Engineer’s Pocket Reference” e-book *. Read more blogs about precision amplifiers. Learn about TI’s entire portfolio of amplifier ICs and explore technical resources. *This requires a myTI log-in.opampportalOPA333SPICE blogOPA2333pampportalPrecision Op Ampstina-tiBlog Post: How does precision lead to automotive safety?http://e2e.ti.com/blogs_/b/behind_the_wheel/archive/2016/09/30/how-does-precision-lead-to-automotive-safetyFri, 30 Sep 2016 08:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:b914f424-355e-4c12-8395-cd30afa7e232Soufiane BendaoudSemiconductor content keeps increasing in automobiles, thanks to the number of sensing technologies. In a span of ten years, the number of sensors has increased steadily across all sensor types. This trend is likely to continue, as more features, which were previously only offered in luxury vehicles or available for purchase after-market, become crucial and in some cases mandated by governing bodies. Advanced driver-assistance system (ADAS) solutions are one of the fastest-growing automotive sectors; the sector is expected to grow by 10 percent from 2015 to 2020, according to a forecast by Strategy Analytics. Even designers of entry-level models now expect ADAS features. As a result, car manufacturers try to meet demand by implementing these features even in entry-level models. The most popular ADAS applications consist of collision avoidance, lane-departure detection, park assist and adaptive cruise control. Depending on geographical area, some applications may be more desirable than others. For example, in densely populated regions, consumers are more likely to want collision warning in their cars, whereas drivers in mountainous areas may feel the need for dynamic lighting. In cities like Shanghai, Moscow, Mumbai or Istanbul, drivers may want object detection with a high level of accuracy. In crowded cities like these, pedestrians, motorcycles and buses can emerge at any given moment. To assist the driver, the car must be able to respond quickly, based on an accurate measurement of the distance between the car and the object. Object-detection applications usually require precision circuits, such as operational amplifiers (op amps), which serve as the fundamental building blocks of a signal chain or analog front end that control the rest of the circuitry in these driver assistance systems. Precision devices, such as the OPA2320-Q1 op amp that has been used into many ADAS designs, provide a low offset voltage and wide bandwidth, which help eliminate system calibration. Because system calibration can involve complex algorithms or even a power-hungry DSP in some design configurations, designers can save on software costs and design time by using precision analog integrated circuits (ICs). While a low offset voltage is necessary to achieve overall system accuracy, the op amp’s wide bandwidth plays a crucial role for settling time. The op amp should settle within ½ LSB of the data converter it drives in order to maintain adequate signal acquisition and integrity. Figure 1: Settling time of OPA2320-Q1 with ±2V input step at a gain of +1 If you’re designing automotive applications, be sure you’ve subscribed to the Behind the Wheel blog to receive more system-level advice and insight on trends. Additional resources Explore the TI Designs library for reference designs for ADAS applications. Find the right precision op amp for your design http://www.ti.com/lsds/ti/amplifiers/op-amps/precision-op-amps-overview.page. Watch this video for an overview of op amp technology .opampportalAdvanced Driver Assistance Systems (ADAS)precision op amp blogsDrive With UspampportalOPA2320-Q1Precision Op AmpsBlog Post: How to design cost-sensitive DC instrumentation circuitshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/08/19/how-to-design-cost-sensitive-dc-instrumentation-circuitsFri, 19 Aug 2016 14:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:5df1b659-7ccb-440f-95f9-8a77ed058b73Collin WellsMany sensors produce low-level DC outputs that require a high input-impedance amplification stage to increase the signal amplitude. Sensors used in personal and portable electronics require operational amplifier (op amp) circuits that provide high input impedance and DC precision, while also being low power and cost-effective. In this post, I’ll explain how to design a few cost-optimized low-power DC-accurate circuits using TLVx333 op amps in different circuit configurations. These devices provide high levels of DC accuracy with maximum input offset voltages (V OS ) less than 15µV, and a typical V OS drift of 0.02µV/°C. The 0.1Hz to 10Hz low-frequency noise specification is only 1.1µVpp and the 0.01 – 1Hz noise specification is only 0.3 µVpp. Table 1 shows the key performance metrics for TLVx333 family. Table 1 : Key specifications for the TLV333 Single-ended sensors can interface with standard noninverting amplifier circuits, as shown in Figure 1. The transfer function is shown in Equation 1. Noninverting op amp circuit-offset errors are dominated by the input offset voltage (V OS ) and the V OS temperature drift of the op amp. Additional offset errors come from the CMRR and the input bias current of the op amp. The tolerance and temperature coefficient of the resistors in the feedback network set the gain error and gain-error drift. The circuit shown in Figure 1 is configured for a gain of 500V/V, and the closed-loop bandwidth is 1.14kHz. Figure 1: TLV333 used in a noninverting amplifier configuration Sensors with differential outputs such as bridge sensors and strain gauges require a circuit with differential inputs. One of the simplest options to interface with a differential sensor is the four-resistor difference amplifier circuit shown in Figure 2. If R 1 is set equal to R 3 and R 2 is set equal to R 4 , then the transfer function simplifies to Equation 2. The tolerance of the resistors in the difference amplifier will directly affect the CMRR of the circuit. Selecting 0.1% resistors achieves at least 54dB of CMRR, while 0.01% resistors achieve at least 74dB. Note that discrete difference amplifier designs will typically not match the performance of integrated solutions, but they often offer advantages in flexibility and cost. The circuit in Figure 2 is configured for a gain of 499V/V, with a closed-loop bandwidth of 1.16kHz. Figure 2: TLV333 used in a difference amplifier configuration High-impedance sensors with differential outputs often require circuits with input impedances >1MΩ. Achieving input impedances >1MΩ is often not practically possible using a discrete difference amplifier topology. Large resistors will increase the DC errors from input bias current, increase circuit intrinsic noise, increase susceptibility to extrinsic noise and will likely require stability compensation. Figure 3 shows a discrete two-op-amp instrumentation amplifier (INA) using a dual-channel TLV2333 . The two-op-amp INA presents a high-impedance differential input to the sensor while only requiring two op amps and five precision resistors. Assuming that R 1 is set equal to R 3 and R 2 is set equal to R 4 , Equation 3 shows the transfer function. The circuit in Figure 3 is configured for a gain of 500V/V, with a closed-loop bandwidth of 1.02kHz. You can also construct a discrete three-op-amp INA using a dual-channel op amp, a single-channel op amp and seven precision resistors. Equation 4 shows the transfer function for the three-op-amp INA. INA designs often require a buffer for a high-impedance reference or an op amp used as an integrator to high-pass filter the input signal. Figure 4 shows a TLV4333 used to create a three-op-amp INA with a reference buffer. The circuit in Figure 4 is configured for a gain of 500V/V and has a closed-loop bandwidth of 1.16kHz. You can use the TLVx333 family of devices in several ways to create DC-accurate circuits that are ideal for cost-optimized precision-sensor acquisition and precision-instrumentation applications. Have questions about other op-amp designs? Log in and leave a comment. Additional resources Read Pete Semig’s articles on V OUT vs. V CM limitations to avoid common pitfalls when using INAs: “ Instrumentation Amplifier V CM vs. V OUT Plots”: Part 1 , Part 2 , Part 3 . “ V CM vs. V OUT plots for instrumentation amplifiers with two op amps .” See TI’s portfolio of performance op amps for cost-conscious applications . Watch more than 40 on-demand precision amplifier training videos in our TI Precision Labs – Op Amps series. Find commonly used analog design formulas in the Analog Engineer’s Pocket Reference e-book. Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalgeneral purpose op amp blogsTLV4333pampportalTLV333Precision Op AmpsTLV blog seriesTLV2333Blog Post: How to design current sensing and protection with off-the-shelf op ampshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/08/12/how-to-design-current-sensing-and-protection-with-off-the-shelf-op-ampsFri, 12 Aug 2016 20:43:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:5770fd60-2b9b-4fb5-9330-3716bdc18de5Hooman HashemiProtecting expensive, critical or hard-to-repair equipment against overcurrent and power-supply fault conditions can be achieved with universally available operational amplifiers ( op amps ) and a few external components. In this post, I will present one example of a versatile variable load-current detection/protection scheme that you can easily alter for a large range of load currents, as it has ubiquitous op amps at its core. “Variable” refers to the benefit of having a nominal “operating” power-supply load-current limit (I_load) that drops on demand. An example would be if you needed to throttle the main system’s power-supply current limit in a transient power-up condition before one or more other supplies have reached their nominal voltage. In Figure 1, if Vref (which represents another monitored power supply) is below nominal (Vnom), the load current drops for safety or reliability reasons. With Vref less than Vmin, the load current pinches off (I_pinch). Figure 1: Load current is a variable function of another supply voltage (Vref) Once Vref is at Vnom or higher, normal I_load limit is permitted to flow. Figure 2 uses the versatile LM7301 rail-to-rail input and output op amp to implement the load current limit profile shown in Figure 1. The circuit monitors the Vsupply current to the load. When the current exceeds the limit determined by the Vref voltage, it produces an output that turns Q1 on and can trigger a protection mechanism. The op amp’s large operating supply voltage (1.8V to 32V) simplifies the design task by extending the range of usable supply voltages (Vsupply) monitorable for load current. In addition, a full-range output swing eases the output drive (on/off) to the gate of a protection transistor or MOSFET (Q1 in Figure 2). Since the input common-mode voltage range extends from below ground to above V+, you can tie the high-side sense resistor (Rsense) directly to the op amp inputs. U1A, which monitors Vref, must have an output swing close to Vsupply in order to allow the kind of behavior depicted in Figure 1, where a reduced Vref pinches off the allowable load current. Furthermore, you could use the same op amp as an amplifier (U1A) or comparator (U1B) to reduce the bill of materials (BOM). Figure 2: Variable current-limit detector The circuit operates by passing all of the load current through a single sense resistor (Rsense), with U1B monitoring both terminals. Enough load-current flow will cause the U1B output to switch high toward the Vsupply rail, which then could turn on a protection device such as Q1. U1A monitors Vref; its output changes the voltage that appears on the non-inverting input of U1B. A low Vref voltage raises U1A output and reduces the I_load value that triggers the U1B output high (fault condition), and vice versa. Diode D1 turns on when U1A detects Vref approaching Vnom and prevents any further increase in I_load with increasing Vref voltage (see Figure 1, where I_load is maintained for Vref ≥ Vnom). The hysteresis resistor R7 works with other external resistors to set the amplitude of the hysteresis, which introduces a difference between the load current that initiates overcurrent and the load current that resets overcurrent. This difference in currents ensures that the circuit does not enter an unstable condition where the U1B output chatters back and forth. At a 4MHz gain-bandwidth product, the op amp can respond to fast current transients if necessary. However, capacitors C1 and C2 can slow down the circuit response time so that transient current spikes do not trigger the overcurrent limit detection – such as those encountered at startup when the supply decoupling capacitors draw excess current to reach their operating voltage. Here are some of the governing equations that make it easier to modify the circuit for different operating conditions. I’ve also included an example operating condition to allow numerical results using the component values shown in Figure 2. Vsupply = 12V Vref = 5V (Vnom condition) To find the current limit as a function of Vref (or U1A out ): When Vref drops, U1A output moves high until U1A output saturates with the values shown. Vmin corresponds to the Vref voltage where the U1A output has saturated high. For a rail-to-rail output device, that means: To find the Vmin in Figure 1: Calculate Vmin with Equation 3 and rearrange Equation 1 to solve for Vref as Vmin: Any lowering of Vref below Vmin has no effect on the load-current limit, which is already pinched off (I_pinch). For the LM7301 , the saturated U1A out voltage is about 100mV lower than Vsupply, or: To find I_pinch in Figure 1, plug the information from Equation 5 into Equation 2: To find the amount of hysteresis in the load current detection point: So lowering the value of R7 increases hysteresis proportionally. Increasing Vref beyond Vnom is clamped by D1 such that the load-current limit remains constant. The value of Vref when this occurs has to do with the voltage divider set by R12 and R13. With the values shown in Figure 2, the D1 anode is set to 8.7V and starts conducting when Vref ≥ 5V, thus establishing Vnom=5V. The voltage divider resistor values should be low enough to supply the current to keep D1 forward-biased with U1A out saturated to ground. Once you have all of the governing expressions for the most important operating points of the circuit, you can easily modify it to fit your intended application. Having a versatile op amp as the main active element in a system can offer added flexibility in setting the operating conditions and load current profile. As an added benefit, it is possible to have more than one supply voltage throttle the load current; just add a series resistor from these other supply voltages to the U1A inverting node, similar to Vref. What considerations do you face when protecting equipment against overcurrent and power-supply fault conditions? Log in to post a comment or visit the TI E2E™ Community Precision Amplifiers forum . Additional resources For more information on dedicated current sense amplifiers, read Dan Harmon’s blog, “ How to get started with current sense amplifiers – part 1 .” Read the Precision Hub blog post, “ Circuit-protection basics .” Start designing with these evaluation modules: Evaluation board for high-speed single op amp in the 5- to 6-pin SOT-23 package . Evaluation board for high-speed single op amp in the 8-pin SOIC package . Universal operational amplifier evaluation module . Download the LM7301 PSPICE model to simulate your designs. Watch the video, “ When to choose a current sense amp .” Search TI high-speed op amps and find technical resources.opampportalsensingpampportalLM7301Blog Post: Go differential to differentiate your precision designhttp://e2e.ti.com/blogs_/b/analogwire/archive/2016/08/11/go-differential-to-differentiate-your-precision-designThu, 11 Aug 2016 15:50:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:d3e84ff8-78bf-440a-8412-d6ee5405e2a0Jason ClarkWhile more of the industry’s newest high-resolution, precision analog-to-digital converters (ADCs) implement differential inputs to maximize performance, many designers still choose to use single-ended amplifiers because that’s what they’re comfortable with. But low-power fully differential amplifiers (FDAs) offer many system advantages without sacrificing precision. In this post, I will use the new THS4551 low noise, precision, 150MHz FDA as an example of how to realize many of the benefits of fully differential amplifiers. FDAs enable simple single-ended to differential signal conversion with direct current (DC) coupling. In Figure 1, you can see three different examples of driving a single-ended signal into the differential input of an ADC. However, the FDA offers lower power, lower noise and an improved dynamic range. (a) (b) (c) Figure 1: Pseudo differential input (a); dual operational amplifier (op amp) method (b); and fully differential amplifier method (c) The FDA architecture can help significantly lower total harmonic distortion (THD) by reducing HD2. Implementing the FDA method shown in Figure 1c, you can achieve an improvement of >4dB in THD. This >4dB improvement can result in an overall system performance improvement, or give you the flexibility to use a lower-power/lower-bandwidth amplifier to meet the same THD. A single FDA (Figure 1c) will have 1/√2 lower noise for the same power than a pair of single-ended op amps (Figure 1b). For example, an op amp with an input-voltage noise of 3nV√Hz will have a total input-voltage noise of 3*√2 nV/√Hz in the dual op amp circuit shown in Figure 1b. The FDA can operate from a single supply voltage and still accept bipolar input signals. Figure 2 shows the THS4551 accepting a 20Vpp input (0V common mode) and outputting 8Vpp with a 2.5V common mode. This capability allows you to reduce system complexity by eliminating the negative power supply and any unnecessary signal-attenuation stages. Figure 2: FDA attenuation example FDAs include a common-mode output loop to perfectly match the expected ADC input common mode. The Vocm pin sets the output common mode of the amplifier. You can leave this pin floating if your desired common mode is at the midpoint of the supplies. As shown in Figure 3, TI FDAs are offered in a number of small packages including 2mm-by-2mm QFN, making them suitable for use in even the most space-constrained application. (a) (b) (c) Figure 3: 2mm-by-2mm 10-pin QFN (a); 3mm-by-3mm 16-pin QFN (b) ; 5mm-by-3mm 8-pin VSSOP (c) TI’s new THS4551 is one of the highest-precision FDAs in the industry, with ±0.175mV input offset and <2V/C offset drift. This enables the improved system performance and minimizes the need for costly and time-consuming system calibrations. The evaluation module (EVM) for TI’s new ADS127L01 24-bit 512Ksps delta-sigma ADC offers an example of the power of FDAs. The EVM for the ADC implements an ADC driver using the THS4551 configured as a multifeedback (MFB) filter. As Figure 4 shows, the ADC-plus-driver pair achieves a signal-to-noise ratio (SNR) of 110.6dB and a THD of 119.1dB with a 1kHz input signal. As shown in the ADS127L01 data sheet (and the goal of all ADC driver implementations), the performance of the THS4551 does not have any impact on the performance of the data converter. This level of performance was achieved while adding less than 7mW of system power, making the THS4551 an essential part of designs that require the lowest power while also delivering the best harmonic distortion and precision. Figure 4: ADS127L01 with THS4551 spectrum If your ADC has a differential input, a precision FDA, such as the THS4551 could be a good choice to simplify your system design and enable low noise, low power, and low harmonic distortion. What is your experience designing with FDAs? Login and leave a comment below about your experience. Additional resources Try an FDA in a system design using one of our TI Designs reference designs, including: Data Acquisition Optimized for Lowest Distortion, Lowest Noise, 18 bit, 1Msps Reference Design (TIPD115). Ultrasonic Water Flow Measurement Reference Design (TIDM-ULTRASONIC-WATER-FLOW-MEASUREMENT) . High Performance Single Ended to Differential Active Interface for High Speed ADC Developed by Dallas Logic Corp. (TIDA-00294). Read more blogs on differentiated amplifiers . Read application notes on FDAs: Using single-supply fully diff. amps with neg. input voltages to drive ADCs Using fully differential op amps as attenuators Analysis of fully differential amplifiers Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalFully Differential Amplifiershsamps_fdaTHS4551TIDA-00294TIPD115smart gridtest measurementTIDM-LC-WATERMTRpampportalADS127L01Precision Op AmpsTIDM-ULTRASONIC-WATER-FLOW-MEASUREMENTfactory automationFully differential amp blogs