Blogshttp://e2e.ti.com/blogs_/Welcome to the Blogs section of the TI E2E Community! Ask questions, share knowledge, explore ideas, and help solve problems with fellow engineers on TI’s Engineer-to-Engineer (E2E) Communityen-USZimbra Community 8Blog Post: “Trust, but verify” SPICE model accuracy, part 3: slew rate and input clamping diodeshttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/18/trust-but-verify-spice-model-accuracy-part-3-slew-rate-and-input-clamping-diodesMon, 18 Sep 2017 17:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:01fde1b8-0655-4eca-9706-071290f0838dIan WilliamsPrevious installments of this blog post series discussed the need to verify SPICE model accuracy and how to measure common-mode rejection ratio (CMRR) and offset voltage versus common-mode voltage (V OS vs. V CM ). In part 3, I’ll continue by explaining how to verify an operational amplifier (op amp) model’s slew rate, which is a large-signal output response. Slew rate Slew rate is defined as the maximum rate of change of an op amp’s output voltage and is typically given in volts per microsecond (V/µs). Slew rate is a type of output distortion, or nonlinearity. An amplifier in this condition is not behaving linearly where the output voltage equals the input voltage multiplied by the closed-loop gain. Instead, the op amp output voltage changes with a constant slope. This continues until the op amp corrects the difference at its input pins and the amplifier returns to a linear or small-signal operating state. For a more detailed look at slew rate, watch our TI Precision Labs – Op Amps video series on slew rate . One of the most common ways to force an op amp into slew rate limit is to apply a large-signal input step of 100mV or greater, but slew limiting can also occur when trying to output high-amplitude signals at high frequencies. In audio applications, for example, slew limiting distorts sine waves into triangle waves, causing visible (and audible) distortion, as shown in Figure 1. Figure 1: Triangular distortion caused by slew rate limit When using SPICE simulation for audio applications or circuits where large-signal input steps are common (such as those with multiplexers or switches), I recommend verifying the slew rate behavior of your op amp model. Figure 2 shows the recommended test circuit. Figure 2: Slew rate test circuit This circuit places the op amp in a unity-gain buffer configuration and applies a large-signal step to the noninverting input pin. The amplitude of the step should match the test conditions given in your specific op amp’s data sheet. Let’s test the slew rate of the OPA196 , a new e-Trim TM precision amplifier from TI, whose data sheet specifies a 10V input step test condition (Figure 3). To determine the slew rate, measure V OUT and calculate its rate of change as it transitions from 10% to 90% of the total output step. Figure 3: Slew rate test results for the OPA196 Equations 1 through 3 calculate the rising slew rate for the OPA196 : where ΔV is the change in voltage and Δt is the change in time from 10% to 90% of the output transition. In this case, the simulated rising slew rate perfectly matches the data sheet spec of 7.5V/µs! You can repeat this test with a negative input step to measure the falling slew rate. Testing amplifiers with input clamping diodes Testing the slew rate of certain types of amplifiers requires a small tweak to the slew rate test circuit. On most bipolar, high-voltage complementary metal-oxide semiconductor (CMOS) and chopper amplifiers, clamping diodes are present across the op amp input pins. If you try to apply a large-signal step to these devices, the large differential input voltage will cause these diodes to conduct current directly from the noninverting input to the inverting input and output. The result is an incorrect slew rate measurement that’s faster than what the actual device can generate. Figure 4 shows this effect with the OPA1678 , a high-voltage CMOS audio amplifier from TI. Figure 4: Slew rate test results (no input current limit) for the OPA1678 This effect was not evident on the OPA196 , even though it’s a high-voltage CMOS amplifier. That’s because it’s part of TI’s OPA19x family of amplifiers with multiplexer-friendly inputs. The design of the OPA19x family eliminates the need for input clamping diodes, and the amplifiers can handle large differential input voltages without them. Junction field-effect transistor (JFET)-input amplifiers such as the OPA145 also do not exhibit this issue. To test the true slew rate of amplifiers with input clamping diodes, place a current-limiting resistor either in series with the input source or between the inverting input and the output pin. A resistance of 10kΩ does the trick for the vast majority of op amps. Figures 5 and 6 show the modified circuit and test results. Figure 5: Slew rate test circuit with input current limit Figure 6: Slew rate test results with input current limit for the OPA1678 Using Equations 1-3 once more, the rising slew rate of the OPA1678 model is 8.9V/µs – very close to the data sheet spec of 9V/µs. As an alternative to using a large input current-limiting resistor, you can test the slew rate with the amplifier in an inverting configuration. In this case, the input and feedback resistors limit the input current through the input diodes and enable an accurate measurement. Thanks for reading the third installment of the “Trust, but verify” blog series! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Read the Bruce Trump article, “ Slew Rate – the op amp speed limit .” To learn more about op amp input clamping diodes, watch the video, “ TI Precision Labs – Op Amps: Comparator Applications 4 .” Download the TI reference guide, “ TI Precision Designs: Reference Design Single Op Amp Slew Rate Limiter .” SaveopampportalOPA1678SPICE blogaudioopampportalOPA145pampportalaudioportalPrecision Op AmpsOPA196Blog Post: Speed up basic circuit design with the analog engineer’s calculatorhttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/07/speed-up-basic-circuit-design-with-the-analog-engineer-s-calculatorThu, 07 Sep 2017 15:30:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:4f7e1a6c-6df6-44a0-901e-bc68de7a9689Jim CattQuick quiz: Can you find the standard 1% resistor values for a voltage divider that comes closest to a divider ratio of V OUT /V IN = .3278, with less than 0.01% error? The answer is 324Ω and 158Ω, with 0.00025% error. Setting up the equation to solve for one of the values in terms of the other is easy. But iterating through multiple standard resistor values is tedious and time consuming, even when using a spreadsheet. The analog engineer’s calculator simplifies this task. This newly developed tool is a companion to the “ Analog Engineer’s Pocket Reference .” Many of you are familiar with this e-book, which covers many fundamental topics in circuit design: unit conversion, components, circuit equations, op amps, printed circuit board (PCB) design, sensors and analog-to-digital converters (ADCs). For those of you who hate memorizing even basic formulas and equations (or more likely, have gotten a little rusty), the pocket reference is an easily accessible source that can save tons of time (unless, of course, you have meticulously indexed your college textbooks). This beta tool contains a collection of simple-to-use calculators that support much of the content in the pocket reference. While it doesn’t address every topic, it does cover the more interesting and complex topics, and constitutes one-stop shopping for many of the simple calculations that you might perform regularly. Figure 1 lists the possible calculations. Figure 1 : Analog engineer’s calculator menu The calculator is especially useful when designing sensor signal-conditioning and data-acquisition systems to monitor voltage, current and temperature. The built-in calculators for amplifiers, data converters and temperature sensors make the task easier and faster. Need to design an input drive circuit for a successive approximation register analog-to-digital converter (SAR ADC)? Use the ADC SAR drive calculator to design the circuit. As Figure 2 shows, simply select the input type (single ended, differential, etc.); enter the ADC resolution, sampling cap value, full-scale input range and acquisition time; and click OK to see the associated resistor-capacitor circuit values, as well as other parameters. Figure 2 : ADC SAR drive calculator Analog designers often need to make cascaded noise calculations when selecting circuit components to meet target specifications. Setting up signal-chain noise calculations can be tedious, but the calculator enables quick computations using only a few input parameters, as shown in Figure 3. Figure 3 : ADC plus signal-chain noise calculation How many times have you designed a simple inverting or noninverting gain stage and wanted to get as close as possible to your target gain using only standard 1% resistors? You probably know that breaking up the feedback resistance into two or more discrete values reduces the error caused by the resistor tolerances. Selecting the 1% resistor values and calculating the actual gain and gain error isn’t difficult, just tedious. You have more important things to do. Use the amplifier gain resistor calculator to speed up the task (Figure 4). Figure 4 : Gain resistor calculator Perhaps you need to calculate the inductance and capacitance of a section of PCB trace. Use the microstrip calculator to quickly find these values by entering a few trace parameters (Figure 5). Figure 5 : Microstrip calculator These examples represent just a few of the often-used analog design calculations, aggregated into a single tool that you can place on your desktop and access offline. No more bouncing between bookmarks on the web. Download the “ Analog Engineer’s Pocket Reference ” and test out the analog engineer’s calculator to begin exploring the useful features that will speed up basic design tasks and save you valuable time. Sign in and comment below if you have any feedback or suggestions about the calculator. Additional resources Learn more about TI’s amplifier and data converter portfolios.opampportalADC blogspadcportalprecision op amp blogspampportaldcportalSAR ADC blogsBlog Post: “Trust, but verify” SPICE model accuracy, part 2: input offset voltage vs. input common-mode voltagehttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/09/05/trust-but-verify-spice-model-accuracy-part-2-input-offset-voltage-vs-input-common-mode-voltageTue, 05 Sep 2017 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:8bd27877-ca3b-47f5-924f-eda542545bc1Ian WilliamsIt’s no secret that low-voltage rail-to-rail input operational amplifiers (op amps) are gradually taking the place of traditional high-voltage amplifiers in many precision applications. Rail-to-rail input amplifiers are extremely useful, since their linear input-voltage range spans the entire power-supply voltage range (or even beyond). They traditionally achieve this span through the use of two pairs of input transistors instead of one pair, but you should be mindful of new design challenges that this topology creates. One challenge is the change in the op amp’s input offset voltage (V OS ) when the amplifier input stage crosses over from one pair of transistors to another. This phenomenon is often called input crossover distortion. V OS is an important performance characteristic of a precision op amp, and many systems must calibrate out the initial offset voltage to meet their performance goals. Any changes to V OS , whether caused by changes in input common-mode voltage (V CM ), temperature or other variables, are highly undesirable and can throw off a system’s total error performance. Figure 1 gives an example of V OS changing dramatically with increased V CM . Figure 1: V OS vs. V CM When using SPICE simulation for rail-to-rail input amplifier designs, it’s wise to check that the V OS vs. V CM behavior of your models matches the real devices. Figure 2 shows the recommended test circuit. Figure 2: V OS vs. V CM test circuit This simple circuit places the op amp in a unity-gain buffer configuration to prevent output swing limitation issues, then sweeps V CM to determine the change in V OS . To plot V OS vs. V CM , run a DC transfer characteristic while stepping V CM across the entire supply voltage range and measure V OS across the op amp input pins as shown in Figure 2. Let’s use this method to test the response of the OPA388 , a new zero-crossover precision amplifier from TI that uses a charge pump in its input stage to achieve true rail-to-rail performance using only a single transistor pair. This eliminates the input crossover distortion found in traditional rail-to-rail input op amps. See Figure 3. Figure 3: V OS vs. V CM results of the OPA388 The simulated results match the responses of the three test devices given in the OPA388 data sheet very closely, with a change of less than 1μV over the entire V CM range. Let’s use the same test circuit to check the response of the OPA2325 , another zero-crossover precision amplifier from TI. See Figure 4. Figure 4: V OS vs. V CM results for the OPA2325 Again, the simulated results match the real silicon very well. Keep in mind that while the simulation model looks like it has higher offset than the real silicon, all of the test devices measured in this plot had a V OS lower than the typical spec of 40μV, while the SPICE model was designed to match the typical. Thanks for reading the second installment of the “Trust, but verify” blog series! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the TI E2E™ Community Simulation Models forum . Additional resources Download the TI tech note, “ Zero-Crossover Amplifiers: Features and Benefits .” Watch these videos to learn more about V OS vs. V CM : “ TI Precision Labs – Op Amps: Low Distortion Design 2 .” “ TI Precision Labs – Op Amps: Input and Output Limitations 1 .”OPA2325opampportalOPA388SPICE blogpampportalPrecision Op AmpsBlog Post: Transimpedance amplifier designs for high-performance, cost-sensitive smoke detector applicationshttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/08/10/transimpedance-amplifier-designs-for-high-performance-cost-sensitive-smoke-detector-applicationsThu, 10 Aug 2017 14:37:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:58ef0f76-6fa0-4b21-9ed2-4f19253d6531Amanda WeiseThis post is co-authored by Collin Wells . Photodiode-based light sensing is a common technique where operational amplifiers (op amps) are used to condition a photodiode sensor for a wide range of applications. An example of this is in smoke detectors, where a photoelectric smoke alarm is used to identify the presence of smoke in the chamber. A photodiode sensor produces a current proportional to the light level presented to it. Depending on the application, the photodiode is operated in either a photovoltaic or photoconductive mode; each has its own merits, which Bruce Trump discussed in detail in this post from his blog, The Signal . In a smoke detector system, the photodiode operates in a photoconductive mode, meaning you will typically use a transimpedance amplifier to amplify the photodiode current. In photoconductive mode, the photodiode is held at a zero-volt (Figure 1a) or reverse voltage bias (Figure 1b), preventing it from forward biasing. Figure 1: Photodiode in photoconductive mode with zero-volt bias (a); or reverse-voltage bias (b) Equation 1 calculates the direct current (DC) transfer function for the circuits shown in Figure 1 (note that the photodiode current (i D ) is flowing away from the op-amp inverting node): The three-step process outlined in John Caldwell’s series on transimpedance amplifiers (see part 3, “ What op amp bandwidth do I need ?”) determines the minimum required op-amp gain bandwidth for transimpedance configurations. The minimum bandwidth is based on the required transimpedance gain and signal bandwidth, along with the total capacitance presented to the inverting node of the op amp. The diode capacitance often dominates the inverting-node capacitance, but don’t forget to include the effects of the op-amp input capacitance. We summarized the three steps explained in John’s posts here for quick reference. 1. Choose the maximum feedback capacitance (C F ) based on the feedback resistor (R F ) and the signal -3dB bandwidth (f P ) (Equation 2): 2. Calculate the total capacitance (C IN ) at the inverting input of the amplifier. For the circuits shown in Figure 1, this is equal to Equation 3: where C J is the diode junction capacitance, C D is the op-amp differential input capacitance and C CM2 is the op-amp inverting input common-mode input capacitance. 3. Calculate the minimum required op-amp gain bandwidth product (f GBW ) (Equation 4): By following these three simple steps, you can avoid many of the stability and performance issues commonly associated with transimpedance amplifier circuits by selecting an amplifier with sufficient bandwidth to perform the required transimpedance gain at the desired signal bandwidth. Op amps with GBWs between 1 - 20MHz are well suited for smoke detector applications because they are able to amplify the low-level signals in the system to sufficient levels while maintaining stability. Along with meeting bandwidth requirements, the op amp must also meet the system’s DC accuracy requirements. The most important DC specification in many transimpedance applications is the input bias current (i B ) of the op amp. i B will directly sum or subtract with the input signal current, which can cause large errors depending on the magnitude of i B compared to the signal current. In smoke detector applications, this DC voltage enables system designers to set the thresholds for the amount of smoke detected before an action is taken in the system. The example shown in Figure 2 uses a 5MΩ resistor to apply a 5MV/A gain to a 100nA full-scale input current. With the input bias current set to 0A, the full-scale output voltage is 500mV – which is expected based on the transfer function in Equation 1. The circuit on the right in Figure 2 displays the effects of the same circuit with an op amp i B of 10nA. In this case, the output voltage is 450mV, which shows that the 10nA input bias current caused a 50mV (or 10%) error from the ideal 500mV output signal. Figure 2: Input bias current effects in transimpedance amplifier circuits Equation 5 calculates the percentage error of the full-scale range (%FSR) based on the full-scale input current (i IN_FS ) and the op amp’s i B : The TLV6001 device is part of a family of high-performance general-purpose amplifiers for a wide variety of cost-conscious transimpedance applications, such a smoke detectors. This is due to the op amps balance between GBW (1 MHz), low Iq (100 µA), low input bias current (1 pA) and low input capacitance (6pF). Other key features that make this family attractive for system designers in smoke detectors are the RRIO swings and the EMI hardened inputs that help reject interference from extrinsic noise sources. Table 1 lists different transimpedance gain and bandwidth combinations for the TLV6001 based on Equations 1 through 4. Be sure to keep the total input capacitance below the maximum input capacitance (C IN­_MAX ) to avoid stability issues. Table 1 : Quick design calculator for TLV6001 transimpedance applications Figure 3 shows the simulated step-response results for a 1MV/A gain and 50kHz bandwidth with the maximum 54pF of input capacitance from the photodiode. The output overshoot and ringing are minimal, indicating a stable design. Figure 3: TLV6001 step-response results; gain = 1MV/A, bandwidth = 50kHz Smoke detector applications use op amps in the transimpedance configuration to amplify low-level photodiode currents. Designing the transimpedance circuit for smoke detectors can be simplified to a few easy steps. First, follow the three steps from John’s blog posts to select the required op-amp bandwidth. Then sort the remaining results to find a device with an i B specification that meets the system’s DC requirements. The TLV600x devices highlighted in this blog are a great family of products to design with for cost-sensitive transimpedance applications. Do you have questions about op amp designs in smoke detectors? Log in and leave a comment below letting us know your experience with transimpedance configurations or any questions you have. Additional resources Find commonly used analog design formulas in the “ Analog Engineer’s Pocket Reference ” e-book. Download The Signal e-book for 32 op amp design lessons that will make you a better engineer. See TI’s portfolio of performance op amps for cost-conscious applications. Read the other two parts in John Caldwell’s three-part blog series on transimpedance circuits: “ Part I – What op amp bandwidth do I need? (Transimpedance Amplifiers) .” “ Part II – What op amp bandwidth do I need? (Transimpedance Amplifiers) .”opampportalgeneral purpose op amp blogsTLV6001General Purpose Amplifiergeneral-purpose op ampspampportalprecision op ampPrecision Op AmpsSmoke and Heat DetectorBlog Post: “Trust, but verify” SPICE model accuracy, part 1: common-mode rejection ratiohttp://e2e.ti.com/blogs_/b/analogwire/archive/2017/07/27/trust-but-verify-spice-model-accuracy-part-1-common-mode-rejection-ratio-cmrrThu, 27 Jul 2017 15:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:384ba23f-a49f-4a4a-84b5-9fe7bfc65aceIan WilliamsSPICE simulation is an immensely valuable tool that allows engineers to have high confidence in their analog designs before ever stepping foot in a lab. While many of us live and die by our simulation results, have you ever stopped to question if your favorite operational amplifier’s (op amp) SPICE model matches the specs promised by its data sheet? Our competitors’ models can often be inaccurate or oversimplified, as shown in the closed-loop output impedance (Zout) comparison in Figure 1. In this case, it’s easy to see the difference between what’s promised and what’s measured. This difference can have a big impact on your designs, potentially causing some unwanted surprises when you fire up your circuit in the lab for the first time. Figure 1: Competitor SPICE model Zout comparison TI is proud to have some of the most accurate SPICE models in the industry. That being said, it’s still a good practice to “trust, but verify” that the models used in your circuit are up to the task. In this blog series, I will show you the recommended test circuits for many of the most critical op amp parameters, starting with common-mode rejection ratio (CMRR). An op amp’s CMRR is formally defined as the ratio of its common-mode gain to its differential-mode gain. In practical terms, the CMRR spec tells you how much additional offset voltage is generated at the op amp’s input when the input common-mode voltage changes. A high CMRR is desirable, meaning that you’ll see less additional offset voltage. CMRR also changes over frequency, so TI data sheets provide a curve representing typical CMRR over frequency, as shown in Figure 2. For a deeper discussion of CMRR and its impact on circuit performance, watch our TI Precision Labs – Op Amps video on CMRR . Figure 2: OPA2187 CMRR vs. frequency I’ll provide two different approaches for measuring op amp CMRR in simulation. Both of these methods were developed by my colleague Zak Kaye , who is well on his way to becoming an analog guru! Modified differential amplifier method The first method is a modification of the classic differential amplifier circuit, shown in Figure 3, that will test the CMRR of the OPA2187 , a new chopper amplifier from TI with 140dB of common-mode rejection. Figure 3: CMRR test circuit (modified differential amplifier) This circuit takes advantage of a SPICE simulator’s ability to isolate nodes using voltage-controlled voltage sources. We’ve strategically placed these sources at the op amp input and output to isolate the feedback network from the op amp’s input and output impedance, allowing for better extraction of the CMRR response. To plot the CMRR, simply run an AC transfer function and use post-processing to create a curve for (Vcm / Vos). Figure 4 shows the results for this configuration compared to the OPA2187 data sheet. Figure 4: Modified differential amplifier results The circuit extracted CMRR fairly well, but the simulation doesn’t match the data sheet that closely at high frequencies. Is that because the model is inaccurate, or are there limitations with the test circuit? Let’s try another circuit configuration and find out. Open-loop method The second test method, shown in Figure 5, uses an open-loop approach. Impossibly large inductors create closed-loop feedback at DC while still allowing for open-loop AC analysis, with two instances of the op amp separately measuring the common- and differential-mode gain. Figure 5: CMRR test circuit (open-loop method) To plot CMRR, run an AC transfer function and use post-processing to create a curve for (ADM / ACM). Figure 6 gives the results for this configuration compared to the OPA2187 data sheet. Figure 6: Open-loop method results As you can see, the open-loop results match the data sheet very closely due to the true isolation of the common- and differential-mode gains of the op amp. I recommend this approach whenever possible. INA1650 CMRR measurement Finally, I’d like to show an easy way to measure CMRR on the INA1650 , our new high common-mode rejection audio line receiver. Because of the highly integrated design of the INA1650 , you can simply connect each input to the same AC source, run an AC transfer function, and invert the output in order to extract CMRR, as shown in Figures 7 and 8. Figure 7: INA1650 CMRR test circuit Figure 8: INA1650 CMRR test results Thanks for reading this first installment of the “trust, but verify” blog series! In the next installment, I’ll discuss how to measure open-loop output impedance and small-signal step response to perform stability analysis. If you have any questions about simulation verification, log in and leave a comment, or visit the E2E™ Simulation Models forum . Additional resources Download these TI TechNotes to learn more about TI’s op amps: “ Zero-Drift Amplifiers: Features and Benefits .” “ Offset Correction Methods: Laser Trim, e-Trim™ and Chopper .”opampportalINA1650SPICE blogOPA2187pampportalPrecision Op AmpsBlog Post: Zero out your system error with zero drift, zero crossover and zero hasslehttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/04/21/zero-out-your-system-error-with-zero-drift-zero-crossover-and-zero-hassleFri, 21 Apr 2017 15:30:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:ee0d4a8c-2fa8-49fa-bf3e-3439e31f4571TamaraThis post is co-authored by Richard Barthel and Errol Leon . In applications such as position sensors, data-acquisition systems and resistance temperature detectors (RTDs), it is important to design with high precision in mind. In many cases, designing with precision integrated circuits (ICs) reduces signal-chain complexity, lowers the external component count, and minimizes board space and bill of materials (BOM) costs. The inaccuracies of one device may propagate through with the inaccuracies of another device, resulting in undesirable and unpredictable errors. In the case of a buffer-configured operational amplifier (op amp) at the output of a digital-to-analog converter (DAC), it’s crucial that your DAC and your op amp are precision devices for an accurate output. A traditional rail-to-rail complementary metal-oxide semiconductor (CMOS) amplifier architecture includes two differential pairs, PMOS (blue) and NMOS (red), shown in Figure 1. Together, these two transistor pairs span the entire input common-mode voltage range. When one transistor pair takes over from the other, however, a unique and nonlinear phenomenon known as “input crossover distortion” occurs due to the intrinsic input offset voltage of each of the two input differential pairs, shown in Figure 2. Figure 1: Traditional rail-to-rail CMOS amplifier architecture Figure 2: Input offset voltage vs. common-mode voltage When you connect a traditional rail-to-rail CMOS op amp at the output of a high-precision DAC, the crossover distortion will introduce an error and result in a drastic increase in integral nonlinearity (INL). This may cause the signal to deviate several least significant bits (LSBs) from its ideal value. Now, what does 1LSB mean? Equation 1 is a simple equation to calculate LSB: where N is the DAC’s number of bits. The DAC8830 is a 16-bit DAC. If the voltage reference is V ref = 5V, then: So to deviate more than 1LSB means that you can have more than 76.3µV of error at your output. This can be detrimental to many precision applications, like critical systems where failure has the potential to negatively impact customers’ end products. So how do you fix this? Enter zero crossover! You can span the entire input common-mode voltage range by using a zero-crossover op amp such as the OPA388 . The zero-crossover topology uses an internal regulated voltage charge pump to increase the positive supply voltage and thus achieve linear operation with input common-mode voltages all the way to its rails with a single input transistor pair, shown in Figure 3. This results in true rail-to-rail input operation without a crossover region, and thus no crossover distortion. If you were to connect this kind of op amp at the output of a DAC, the op amp does not introduce an error within the common-mode region (1V to 2V below the positive rail) like a traditional rail-to-rail CMOS device. Figure 3: Zero-crossover amplifier architecture In Figure 4, the black curve describes the output of a traditional rail-to-rail CMOS op amp ( OPA340 ) at the output of a DAC ( DAC8830 ), while the red curve describes the output of a zero-crossover op amp ( OPA388 ) with the same DAC8830 . As you can see, the output of the DAC8830 + OPA388 does not suffer from the distortion that is easily visible in the DAC8830 + OPA340 output curve. The High-Precision Reference Design for Buffering a DAC Signal describes this output in greater detail. Figure 4: INL comparison of rail-to-rail CMOS OPA340 and zero-crossover OPA388 Let’s put this reference design into perspective and use it in an application such as an MRI machine. An MRI uses a powerful magnetic field to produce detailed 2-D and 3-D pictures of the human body to diagnose and/or monitor several health conditions. Unacceptably distorted signals that exceed the error budget in any way can potentially impair the quality of the images. The OPA388 is the industry’s first op amp to employ zero-crossover and zero-drift technology. Zero-drift op amps have an internal self-correcting circuit that produces ultra-low input offset voltage (V OS ) and near-zero input offset voltage drift over time and temperature (dV OS /dT). The technology also delivers other advantages, including no 1/f noise (flicker noise), low broadband noise (white noise) and low output distortion, which can help increase system reliability in harsh environments. Take a swimming pool for instance – pH pool testers and monitoring systems must withstand changes in the environment’s temperature to correctly determine the deficit or excess of chlorine. Since most pools are placed outside, the environment’s temperature can vary many degrees between a cold winter’s day and a hot summer’s day. Offset voltage will change with temperature deviations, introducing error, so it is crucial to select an op amp with low offset voltage drift to support system reliability through these changes. To assure high performance, high precision and high accuracy, carefully select parts for your design. Make sure that you understand your system and what you can afford in terms of error, and only then sift through TI’s diverse portfolio for your ultimate solution. Additional resources Download the zero-drift and zero-crossover Tech Notes. Visit the OPA388 and DAC8830 datasheets to learn more.opampportalprecision op amp blogsOPA388DAC8830pampportalPrecision Op Ampsopa340Blog Post: Threading the needle: new op amp delivers exceptional precision, accuracyhttp://e2e.ti.com/blogs_/b/thinkinnovate/archive/2017/03/06/threading-the-needle-new-op-amp-delivers-exceptional-precision-accuracyMon, 06 Mar 2017 14:29:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:36d7459d-25b8-4463-b849-a65078f9f43fThink. Innovate. contributorPrecision affects every part of our lives – from ordering lunch meat at the deli to calibrating delicate lab equipment. And a new high-precision operational amplifier – a circuit that magnifies tiny electric currents so that they can be measured and managed – offers an exceptional level of accuracy in industrial, automotive, medical, personal electronics, and test-and-measurement equipment. “The performance on this device is so precise that it’s like measuring a dime on the Empire State Building,” said Richard Barthel, a systems engineer who worked on the team that developed the OPA388 . “It takes something very small and gets it right on target.” Operational amplifiers – or op amps – are links between sensors that measure analog signals such as pressure, temperature and flow and the digital brains behind technologies that are so integral to our everyday lives. Sensors pick up analog signals from the environment around us that in many cases are measured in the millionths of a volt – too small to be useful for the circuits that convert them to digital signals. The job of op amps is to boost those signals to higher voltages that can then be measured, interpreted and managed by computers. In that amplification process, any variation in the signal gets progressively more distorted as it works its way through the subsequent signal chain. That, in turn, affects the precision of the final measurements produced by a piece of equipment, said Ying Zhou, product marketing manager for the device. Our zero-drift and zero-crossover technologies – which our newest op amp combines in one device for the first time – correct for any noise and errors in the signals and remove the need for designers to add discrete calibration circuits to the systems they create. Combining these technologies will lead to improvements in the accuracy of measurements in applications ranging from electronic scales to heart-rate monitors and pressure sensors. For example, the device’s precision is beneficial for equipment such as: Gauges used in CT scan machines . This medical equipment requires a smooth and consistent movement so the weight and weight distribution of patients can be measured precisely, a critical factor in accurate diagnosis and treatment. Construction equipment . Contractors, civil engineers or other workers will be able to measure elevation and distance with pinpoint precision during building construction, which could increase structural integrity. Weigh scales. Achieve more precise weight measurements – whether you’re ordering a few slices of lunch meat at a deli or checking the weight of a fully loaded semi-truck. Medical lab equipment. Smaller equipment and more precise diagnoses could mean less time in the clinic and more accurate diagnoses for patients. “Precision affects everybody,” Richard said. “The OPA388 takes very small measurements and gets them right on target with high accuracy and resolution. It threads the needle.” If you are interested in learning more, read these TI Tech Notes regarding the features and benefits of zero-drift and zero-crossover technologies.opampportalprecision op amp blogsOPA388test measurementpampportalPrecision Op AmpsBlog Post: Designing a discrete wide-bandwidth, cost-sensitive instrumentation amplifierhttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/02/03/designing-a-discrete-wide-bandwidth-cost-sensitive-instrumentation-amplifierFri, 03 Feb 2017 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:b40acc93-226b-4961-a180-706b30889dfdCole MaciasIn this post, I’ll show how to design a cost-optimized, discrete, wide-bandwidth instrumentation amplifier using the TLV3544 . Instrumentation amplifiers are used for their high input impedance and ability to convert differential voltages to single-ended voltages. Fast current sensing, precision data acquisition, vibration analysis, microphone pre-amplification, ADC drivers and medical instrumentation are all applications that need instrumentation amplifiers with wide bandwidth. TLV354x devices provide a rail-to-rail input/output, 200MHz unity gain bandwidth (GBW) and 150V/µs slew rate, which is designed for the applications I just mentioned. Figure 1 shows the standard three- operational amplifier (op amp) topology. Figure 1: Discrete three-op-amp topology using the TLV3544 The input stage uses dual noninverting amplifiers that enable high impedance at both inputs, whose gain is defined by RF1 = RF2 and RG1. The output stage consists of a difference amplifier with a low impedance output, whose gain is set by R2 = R4 and R1 = R3. The reference voltage, input stage gain and output stage gain define the output voltage, shown in Equation 1: Note that the tolerance of the resistors in the instrumentation amplifier will negatively affect the CMRR and gain error of the circuit. That is why there is a cost and performance trade-off between discrete and integrated instrumentation amplifiers. The bandwidth of instrumentation amplifiers is bounded by three characteristics: the open-loop gain (Aol) of the op amp, the noise gain (Gn) and filtering. Both Aol and Gn are covered in the TI Precision Labs training series on bandwidth (see part 3, “ TI Precision Labs – Op Amps: Bandwidth 3 ” – viewing requires a myTI login). The output-stage difference amplifier’s noise gain determines the circuit’s bandwidth. The TLV3544 , like many high-speed amplifiers, will have stability issues if the feedback resistors are too large. Figure 2 simulates the effects of 200Ω feedback resistors versus 500Ω feedback resistors. Note that 45 degrees of phase margin (PM) is necessary for stable operation. Increasing the resistor values allows for lower power consumption and larger RG1 values to set the gain. Figure 2: TINA-TI ™ software frequency response showing AC peaking, PM and bandwidth with 200Ω (54 degrees PM) versus 500Ω (26 degrees PM) feedback resistors Since the instrumentation amplifier is discrete, you have access to the feedback paths, which allows you to compensate for AC peaking. By placing capacitors in the feedback path, you introduce multiple poles that create a low-pass filter and attenuate the peaking. Equation 2 calculates the -3dB frequency (f p ) of this low-pass filter: Figure 3 shows the schematic and frequency response. Remember that adding a filter will affect the overall bandwidth of the circuit. Figure 3: TINA-TI software schematic and frequency response of compensated feedback paths Since this instrumentation amplifier topology only requires three op amps, the fourth op amp provided by the TLV3544 can serve as a reference buffer for single-supply systems or as an integrator for high-pass filtering of the input. Thus, you can use the TLV3544 to make a discrete wide-bandwidth instrumentation amplifier that is ideal for cost-optimized, high-precision, wide-bandwidth applications. Have questions or comments about other design considerations for discrete instrumentation amplifiers? Log in and leave a comment. Additional resources Watch more than 40 on-demand videos about topics such as bandwidth and stability on TI Precision Labs . Read Pete Semig’s blog series on V OUT vs. V CM limitations, “Instrumentation amplifier V CM vs. V OUT plots: part 1 , part 2 , part 3 ,” to avoid common pitfalls when using instrumentation amplifiers. See TI’s portfolio of performance op amps for cost-conscious applications. Find commonly used analog design formulas in our wildly popular and free Analog Engineer’s Pocket Reference e-book . Read more blogs about precision amplifiers . Learn about TI’s entire portfolio of amplifier ICs and explore technical resources.opampportalInstrumentation AmplifiersTLV3544analog bloggeneral-purpose op ampspampportaltina-tiTLV blog seriesBlog Post: Achieve big board-size reductions with tiny, precision op ampshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2017/01/20/achieve-big-board-size-reductions-with-tiny-precision-op-ampsFri, 20 Jan 2017 16:27:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:ff00257d-26ee-4fe6-aace-c17e39dd8be1Ying ZHOUElectronics such as smartphones, tablets, notebooks and wearable products are becoming more multifunctional, smaller and slimmer. Achieving higher functionality in smaller form factors requires extremely tiny ICs. Many times, different package types help reduce size and solve various design challenges. Take operational amplifiers (op amps), for example; wafer chip-scale packages (WCSP) typically enable the smallest possible footprint for optical modules and wearables, while flat no-lead packages such as quad flat no-lead (QFN) or dual flat no-lead (DFN) inspire differentiated audio functionality in personal electronics where small size, high performance, and easy testing and tuning are essential. Small-outline transistor (SOT) packages like SOT553 are suitable for emerging industrial applications such as field transmitters, which require a wide supply but a small footprint in a user-friendly leaded package. The rapid evolution of mobile devices is driving package technology innovations, but the smallest possible package will always be the size of the die itself. The WCSP is a type of package that can be almost as small as the die, so it’s desirable in applications like medical diagnostics, fitness monitoring and handheld electronic devices. The OPA2376 in WCSP (1.11mm by 2.15mm by 0.625mm) is a device commonly used for signal conditioning purpose in the areas mentioned above. Audio in portable devices is another example where tiny packages are inspired by the consumer needs. Traditionally in the professional audio world, audiophiles like to use op amps in the dual inline package (DIP), which is more DIY-friendly for audio designers. The OPA2134 from the legacy Burr-Brown TM audio portfolio, is such an example with great audio reputation. However, with the increased demand of high-quality audio in portable devices, tiny-package high-performance audio ICs are required in order to enable the high-fidelity sound in a space-constraint design. With the implementation of the DFN package on the OPA1652 (Figure 1), a performance upgrade to the OPA2134 , you will find a high-performance current-to-voltage converter with great sound quality for cost-optimized portable audio equipment, smartphones and gaming motherboards. Figure 2 shows the total harmonics distortion and noise (THD+N) performance of the OPA1652 and OPA2134 with common-mode impedance mismatching. Figure 1: OPA1652 (DFN) versus OPA2134 (PDIP) Figure 2: THD+N test in a gain of 100: the OPA1652 vs. the OPA2134 A tiny package may not be the only consideration for wearable devices. These types of applications typically spend most of their time in sleep mode until they are needed to measure biometric data. This requirement has inspired op amps with a shutdown feature in tiny QFN packages. For applications where power consumption is a vital concern, tiny packaging helps enable product innovation, because you can achieve both low power and a small form factor. The OPA2316S in a X2QFN (1.5mm by 2mm by 0.4mm) operates down to 1.8V with a wide bandwidth of 10MHz, as well as rail-to-rail performance, making it suitable for battery-powered designs. Some industrial factory automation applications also require a tiny package. Often, designers prefer devices with small-outline lead packages that are easy to prototype, layout and swap with pin-to-pin-compatible ICs. The OPA171 is one of the first micropower 36V op amps offered in both a single SOT553 (1.6mm x 1.6 mm) package and a dual, very-thin shrink small outline package (VSSOP) (2.0mm x 3.1mm), providing an optimized combination of low cost and performance for applications such as tracking amplifiers in power modules, transducer amplifiers and battery-powered instruments.As a fundamental building block for a signal chain, op amps must keep pace with the emerging electronic design trends. TI has developed products in some of the smallest packages available, including WCSP, QFN/DFN and SOT553. With innovations in design and technologies, more tiny-package, low-power precision op amps are on the way. Browse TI’s tiny op amp portfolio to find the best one for your next project. Additional resources Read the Analog Applications Journal article, “ Distortion and source impedance in JFET-input op amps .” Learn more about the TI Design, “ A High-Fidelity Headphone Amplifier for Current Output Audio DACs Reference Design ” Visit the “ Headphone Amplifier for Voltage-Output Audio DACs Reference Design ”opampportalOPA1652OPA171precision op amp blogstest measurementwearablespampportalOPA2134Precision Op AmpsOPA2376Blog Post: Multiplexers: not so simplehttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/11/18/multiplexers-not-so-simpleFri, 18 Nov 2016 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:00783dc3-148c-4bb7-8026-d9913171ee82Evan SawyerIt’s simple to design a multiplexer (or mux for short) into a signal chain, right? After all, the device simply funnels multiple signals into a data converter. In reality, a mux can significantly impact the performance of a signal chain in a variety of ways. For example, the on-capacitance can cause crosstalk between channels. Signal- and temperature-dependent variations in the on-resistance can introduce signal distortion. Together, the capacitance and resistance of the mux can limit signal bandwidth. Charge injection can introduce transient errors when the mux switches channels and impact settling time at the output. To optimize signal-chain performance, it is important to understand these examples as well as numerous other ways that a mux can impact a signal, especially because multiplexers are optimized for different performance characteristics, and thus for different applications. Figure 1 is an example circuit containing a mux with its output connected to an inverting operation amplifier ( op amp ). Figure 1: A mux connected to an inverting amplifier introduces gain error This circuit is one of many common mux configurations in a signal chain, but as we will discover, this design will lead to significant signal gain error. Assuming that the op amp is ideal (no offset, bias current, input/output limitations, etc.), Equation 1 expresses the signal gain as: Because the MUX36S08 is not an ideal mux and has internal capacitance as well as an on-resistance of 125Ω, Equation 2 expresses the effective gain of the system as: The calculated signal gain in Equation 2 poses a huge problem if the output of the op amp is connected to a data converter designed to receive the full gain, as nearly 40% of the converter’s range would not be utilized. This equation doesn’t even take into account the on-resistance variation that occurs from changes in temperature, signal voltage or the voltage applied to the supplies. Figure 2 shows one of the on-resistance curves for the MUX36S08 . You can see that the resistance changes based on the temperature as well as the applied signal (the source or drain voltage). The curve that results from changing the signal voltage is known as on-resistance flatness, which can introduce nonlinearity and gain variation. Subjecting the circuit in Figure 1 to a full ±18V sinusoidal signal and temperatures from -40°C to 125°C, the on-resistance of the mux can vary from approximately 75Ω to 250Ω, resulting in an effective gain ranging from -0.44 to -0.73. Figure 2: On-resistance vs. source or drain voltage Luckily, you can effectively ignore the on-resistance of the multiplexer through very simple design precautions. Figure 3 shows the output of the mux connected to an op amp configured as a buffer. The high input impedance of the op amp eliminates any gain error the system would otherwise experience. Figure 3: A mux connected to a buffer effectively eliminates gain error caused by the mux on-resistance As a reminder, the effect of on-resistance on signal gain is only one of the many ways that a multiplexer can impact the performance of a system. When you’re ready to learn about the other ways a mux can add error and distortion to a signal as well as how to mitigate the impact a mux has on signal chain performance, check out the new TI Precision Labs training series covering multiplexers. Additional resources Explore more than 40 hands-on trainings and lab videos from TI Precision Labs . Read a blog about whether a low-leakage multiplexer really matters in a high-impedance PLC system.opampportalMultiplexer/Demultiplexer (Mux/Demux)MUX36S08pampportalBlog Post: How to fix your simulations when the macromodel’s voltage noise doesn’t match the datasheethttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/09/30/how-to-fix-your-simulations-when-the-macromodels-voltage-noise-doesnt-match-the-datasheetFri, 30 Sep 2016 16:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:217bde76-4214-4ac5-9b85-e93db23bd3e8Cole MaciasWhen responding to questions posted on TI E2E™ Community forums , we frequently run simulations using TINA-TI ™ software, a SPICE-based simulation program. Since we are always in the process of updating our simulation models, we sometimes run across SPICE models that are old, outdated or incorrect when modeling performance parameters. One recent example involves the voltage-noise density of the OPA2333 macromodel. Unfortunately, we found that the model’s voltage-noise density curve was less than that given in the data sheet. So in this blog post, I will show you how to verify an operational amplifier’s (op amp) voltage-noise density curve and correct it if necessary. First, you need to know how to generate a voltage-noise density curve using TINA-TI software. In this example, I will use the OPA2333 macromodel and schematic shown in Figure 1. Figure 1: TINA-TI test bench for voltage-noise density The output noise in this configuration uses the op amp with no gain, filtering or other factors that would change the voltage noise over frequency. To simulate the output noise, select Analysis > Noise Analysis, and tick the Output Noise check box shown in Figure 2. Figure 2: How to find noise analysis for output noise Figure 3: Simulated OPA2333 voltage noise Figure 4: OPA2333 voltage noise according to the data sheet You can add noise to the macromodel by inserting a voltage-noise source in the schematic. To get the voltage-noise source, go to File > Open Examples. Select the Noise Sources folder and open the TINA Noise Sources.TSC file shown in Figure 5. Figure 5: Finding the voltage-noise source and equivalent op amp noise model Now, copy and paste the voltage-noise source from the noise-source schematic into the testing schematic and add it to the noninverting input, as shown in Figure 6. Figure 6: Testing schematic for input-voltage noise with a voltage-noise source Double-click on the noise source and select Enter Macro. A tab will open showing the netlist of the voltage source; see Figure 8. Since the OPA333 is a chopper op amp and has no flicker noise, you do not want to add flicker noise to the voltage source. Find the parameters NLF and FLW and change them to 0 and 0.1, respectively. This sets the flicker noise to 0 nV/ √ Hz at 0.1Hz. You will also need to adjust the broadband noise so that the root sum square (RSS) noise is equal to the value you want. In the netlist, this value is represented by the parameter NVR. In Figure 7, I solve for the desired broadband voltage. Figure 8 shows the updated parameters inside the netlist. Figure 7: Solving for the broadband voltage-noise value Figure 8: Netlist of voltage-noise source Running the noise analysis again (Figure 9), you can see that the voltage noise is now the same value shown in Figure 4. Figure 9: Corrected op amp voltage noise Note that this process only works when your macromodel’s voltage-noise density curve is less than that given in the data sheet because RSS noise can only be added. If your macromodel’s voltage-noise density curve is more than that given in the datasheet, the TI Precision Labs on-demand training series includes a short video on how to create your own accurate macromodel for noise simulations. You can find the video here *. Remember, trust your SPICE models, but verify that they are correct. Simulations can take you far, but they need to simulate correctly to produce meaningful results. Additional resources Learn more about noise , how to calculate it and how to simulate it correctly with our online training series, TI Precision Labs – Op Amps. Learn how to trust and verify your SPICE macromodels in Tim Green’s blog post, “ SPICE op amp macromodels: ‘Trust but verify’ .” Find commonly used analog design formulas in our wildly popular and free “ Analog Engineer’s Pocket Reference” e-book *. Read more blogs about precision amplifiers. Learn about TI’s entire portfolio of amplifier ICs and explore technical resources. *This requires a myTI log-in.opampportalOPA333SPICE blogOPA2333pampportalPrecision Op Ampstina-tiBlog Post: How does precision lead to automotive safety?http://e2e.ti.com/blogs_/b/behind_the_wheel/archive/2016/09/30/how-does-precision-lead-to-automotive-safetyFri, 30 Sep 2016 08:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:b914f424-355e-4c12-8395-cd30afa7e232Soufiane BendaoudSemiconductor content keeps increasing in automobiles, thanks to the number of sensing technologies. In a span of ten years, the number of sensors has increased steadily across all sensor types. This trend is likely to continue, as more features, which were previously only offered in luxury vehicles or available for purchase after-market, become crucial and in some cases mandated by governing bodies. Advanced driver-assistance system (ADAS) solutions are one of the fastest-growing automotive sectors; the sector is expected to grow by 10 percent from 2015 to 2020, according to a forecast by Strategy Analytics. Even designers of entry-level models now expect ADAS features. As a result, car manufacturers try to meet demand by implementing these features even in entry-level models. The most popular ADAS applications consist of collision avoidance, lane-departure detection, park assist and adaptive cruise control. Depending on geographical area, some applications may be more desirable than others. For example, in densely populated regions, consumers are more likely to want collision warning in their cars, whereas drivers in mountainous areas may feel the need for dynamic lighting. In cities like Shanghai, Moscow, Mumbai or Istanbul, drivers may want object detection with a high level of accuracy. In crowded cities like these, pedestrians, motorcycles and buses can emerge at any given moment. To assist the driver, the car must be able to respond quickly, based on an accurate measurement of the distance between the car and the object. Object-detection applications usually require precision circuits, such as operational amplifiers (op amps), which serve as the fundamental building blocks of a signal chain or analog front end that control the rest of the circuitry in these driver assistance systems. Precision devices, such as the OPA2320-Q1 op amp that has been used into many ADAS designs, provide a low offset voltage and wide bandwidth, which help eliminate system calibration. Because system calibration can involve complex algorithms or even a power-hungry DSP in some design configurations, designers can save on software costs and design time by using precision analog integrated circuits (ICs). While a low offset voltage is necessary to achieve overall system accuracy, the op amp’s wide bandwidth plays a crucial role for settling time. The op amp should settle within ½ LSB of the data converter it drives in order to maintain adequate signal acquisition and integrity. Figure 1: Settling time of OPA2320-Q1 with ±2V input step at a gain of +1 If you’re designing automotive applications, be sure you’ve subscribed to the Behind the Wheel blog to receive more system-level advice and insight on trends. Additional resources Explore the TI Designs library for reference designs for ADAS applications. Find the right precision op amp for your design http://www.ti.com/lsds/ti/amplifiers/op-amps/precision-op-amps-overview.page. Watch this video for an overview of op amp technology .opampportalAdvanced Driver Assistance Systems (ADAS)precision op amp blogsDrive With UspampportalOPA2320-Q1Precision Op AmpsBlog Post: How to design cost-sensitive DC instrumentation circuitshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/08/19/how-to-design-cost-sensitive-dc-instrumentation-circuitsFri, 19 Aug 2016 14:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:5df1b659-7ccb-440f-95f9-8a77ed058b73Collin WellsMany sensors produce low-level DC outputs that require a high input-impedance amplification stage to increase the signal amplitude. Sensors used in personal and portable electronics require operational amplifier (op amp) circuits that provide high input impedance and DC precision, while also being low power and cost-effective. In this post, I’ll explain how to design a few cost-optimized low-power DC-accurate circuits using TLVx333 op amps in different circuit configurations. These devices provide high levels of DC accuracy with maximum input offset voltages (V OS ) less than 15µV, and a typical V OS drift of 0.02µV/°C. The 0.1Hz to 10Hz low-frequency noise specification is only 1.1µVpp and the 0.01 – 1Hz noise specification is only 0.3 µVpp. Table 1 shows the key performance metrics for TLVx333 family. Table 1 : Key specifications for the TLV333 Single-ended sensors can interface with standard noninverting amplifier circuits, as shown in Figure 1. The transfer function is shown in Equation 1. Noninverting op amp circuit-offset errors are dominated by the input offset voltage (V OS ) and the V OS temperature drift of the op amp. Additional offset errors come from the CMRR and the input bias current of the op amp. The tolerance and temperature coefficient of the resistors in the feedback network set the gain error and gain-error drift. The circuit shown in Figure 1 is configured for a gain of 500V/V, and the closed-loop bandwidth is 1.14kHz. Figure 1: TLV333 used in a noninverting amplifier configuration Sensors with differential outputs such as bridge sensors and strain gauges require a circuit with differential inputs. One of the simplest options to interface with a differential sensor is the four-resistor difference amplifier circuit shown in Figure 2. If R 1 is set equal to R 3 and R 2 is set equal to R 4 , then the transfer function simplifies to Equation 2. The tolerance of the resistors in the difference amplifier will directly affect the CMRR of the circuit. Selecting 0.1% resistors achieves at least 54dB of CMRR, while 0.01% resistors achieve at least 74dB. Note that discrete difference amplifier designs will typically not match the performance of integrated solutions, but they often offer advantages in flexibility and cost. The circuit in Figure 2 is configured for a gain of 499V/V, with a closed-loop bandwidth of 1.16kHz. Figure 2: TLV333 used in a difference amplifier configuration High-impedance sensors with differential outputs often require circuits with input impedances >1MΩ. Achieving input impedances >1MΩ is often not practically possible using a discrete difference amplifier topology. Large resistors will increase the DC errors from input bias current, increase circuit intrinsic noise, increase susceptibility to extrinsic noise and will likely require stability compensation. Figure 3 shows a discrete two-op-amp instrumentation amplifier (INA) using a dual-channel TLV2333 . The two-op-amp INA presents a high-impedance differential input to the sensor while only requiring two op amps and five precision resistors. Assuming that R 1 is set equal to R 3 and R 2 is set equal to R 4 , Equation 3 shows the transfer function. The circuit in Figure 3 is configured for a gain of 500V/V, with a closed-loop bandwidth of 1.02kHz. You can also construct a discrete three-op-amp INA using a dual-channel op amp, a single-channel op amp and seven precision resistors. Equation 4 shows the transfer function for the three-op-amp INA. INA designs often require a buffer for a high-impedance reference or an op amp used as an integrator to high-pass filter the input signal. Figure 4 shows a TLV4333 used to create a three-op-amp INA with a reference buffer. The circuit in Figure 4 is configured for a gain of 500V/V and has a closed-loop bandwidth of 1.16kHz. You can use the TLVx333 family of devices in several ways to create DC-accurate circuits that are ideal for cost-optimized precision-sensor acquisition and precision-instrumentation applications. Have questions about other op-amp designs? Log in and leave a comment. Additional resources Read Pete Semig’s articles on V OUT vs. V CM limitations to avoid common pitfalls when using INAs: “ Instrumentation Amplifier V CM vs. V OUT Plots”: Part 1 , Part 2 , Part 3 . “ V CM vs. V OUT plots for instrumentation amplifiers with two op amps .” See TI’s portfolio of performance op amps for cost-conscious applications . Watch more than 40 on-demand precision amplifier training videos in our TI Precision Labs – Op Amps series. Find commonly used analog design formulas in the Analog Engineer’s Pocket Reference e-book. Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalgeneral purpose op amp blogsTLV4333pampportalTLV333Precision Op AmpsTLV blog seriesTLV2333Blog Post: How to design current sensing and protection with off-the-shelf op ampshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/08/12/how-to-design-current-sensing-and-protection-with-off-the-shelf-op-ampsFri, 12 Aug 2016 20:43:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:5770fd60-2b9b-4fb5-9330-3716bdc18de5Hooman HashemiProtecting expensive, critical or hard-to-repair equipment against overcurrent and power-supply fault conditions can be achieved with universally available operational amplifiers ( op amps ) and a few external components. In this post, I will present one example of a versatile variable load-current detection/protection scheme that you can easily alter for a large range of load currents, as it has ubiquitous op amps at its core. “Variable” refers to the benefit of having a nominal “operating” power-supply load-current limit (I_load) that drops on demand. An example would be if you needed to throttle the main system’s power-supply current limit in a transient power-up condition before one or more other supplies have reached their nominal voltage. In Figure 1, if Vref (which represents another monitored power supply) is below nominal (Vnom), the load current drops for safety or reliability reasons. With Vref less than Vmin, the load current pinches off (I_pinch). Figure 1: Load current is a variable function of another supply voltage (Vref) Once Vref is at Vnom or higher, normal I_load limit is permitted to flow. Figure 2 uses the versatile LM7301 rail-to-rail input and output op amp to implement the load current limit profile shown in Figure 1. The circuit monitors the Vsupply current to the load. When the current exceeds the limit determined by the Vref voltage, it produces an output that turns Q1 on and can trigger a protection mechanism. The op amp’s large operating supply voltage (1.8V to 32V) simplifies the design task by extending the range of usable supply voltages (Vsupply) monitorable for load current. In addition, a full-range output swing eases the output drive (on/off) to the gate of a protection transistor or MOSFET (Q1 in Figure 2). Since the input common-mode voltage range extends from below ground to above V+, you can tie the high-side sense resistor (Rsense) directly to the op amp inputs. U1A, which monitors Vref, must have an output swing close to Vsupply in order to allow the kind of behavior depicted in Figure 1, where a reduced Vref pinches off the allowable load current. Furthermore, you could use the same op amp as an amplifier (U1A) or comparator (U1B) to reduce the bill of materials (BOM). Figure 2: Variable current-limit detector The circuit operates by passing all of the load current through a single sense resistor (Rsense), with U1B monitoring both terminals. Enough load-current flow will cause the U1B output to switch high toward the Vsupply rail, which then could turn on a protection device such as Q1. U1A monitors Vref; its output changes the voltage that appears on the non-inverting input of U1B. A low Vref voltage raises U1A output and reduces the I_load value that triggers the U1B output high (fault condition), and vice versa. Diode D1 turns on when U1A detects Vref approaching Vnom and prevents any further increase in I_load with increasing Vref voltage (see Figure 1, where I_load is maintained for Vref ≥ Vnom). The hysteresis resistor R7 works with other external resistors to set the amplitude of the hysteresis, which introduces a difference between the load current that initiates overcurrent and the load current that resets overcurrent. This difference in currents ensures that the circuit does not enter an unstable condition where the U1B output chatters back and forth. At a 4MHz gain-bandwidth product, the op amp can respond to fast current transients if necessary. However, capacitors C1 and C2 can slow down the circuit response time so that transient current spikes do not trigger the overcurrent limit detection – such as those encountered at startup when the supply decoupling capacitors draw excess current to reach their operating voltage. Here are some of the governing equations that make it easier to modify the circuit for different operating conditions. I’ve also included an example operating condition to allow numerical results using the component values shown in Figure 2. Vsupply = 12V Vref = 5V (Vnom condition) To find the current limit as a function of Vref (or U1A out ): When Vref drops, U1A output moves high until U1A output saturates with the values shown. Vmin corresponds to the Vref voltage where the U1A output has saturated high. For a rail-to-rail output device, that means: To find the Vmin in Figure 1: Calculate Vmin with Equation 3 and rearrange Equation 1 to solve for Vref as Vmin: Any lowering of Vref below Vmin has no effect on the load-current limit, which is already pinched off (I_pinch). For the LM7301 , the saturated U1A out voltage is about 100mV lower than Vsupply, or: To find I_pinch in Figure 1, plug the information from Equation 5 into Equation 2: To find the amount of hysteresis in the load current detection point: So lowering the value of R7 increases hysteresis proportionally. Increasing Vref beyond Vnom is clamped by D1 such that the load-current limit remains constant. The value of Vref when this occurs has to do with the voltage divider set by R12 and R13. With the values shown in Figure 2, the D1 anode is set to 8.7V and starts conducting when Vref ≥ 5V, thus establishing Vnom=5V. The voltage divider resistor values should be low enough to supply the current to keep D1 forward-biased with U1A out saturated to ground. Once you have all of the governing expressions for the most important operating points of the circuit, you can easily modify it to fit your intended application. Having a versatile op amp as the main active element in a system can offer added flexibility in setting the operating conditions and load current profile. As an added benefit, it is possible to have more than one supply voltage throttle the load current; just add a series resistor from these other supply voltages to the U1A inverting node, similar to Vref. What considerations do you face when protecting equipment against overcurrent and power-supply fault conditions? Log in to post a comment or visit the TI E2E™ Community Precision Amplifiers forum . Additional resources For more information on dedicated current sense amplifiers, read Dan Harmon’s blog, “ How to get started with current sense amplifiers – part 1 .” Read the Precision Hub blog post, “ Circuit-protection basics .” Start designing with these evaluation modules: Evaluation board for high-speed single op amp in the 5- to 6-pin SOT-23 package . Evaluation board for high-speed single op amp in the 8-pin SOIC package . Universal operational amplifier evaluation module . Download the LM7301 PSPICE model to simulate your designs. Watch the video, “ When to choose a current sense amp .” Search TI high-speed op amps and find technical resources.opampportalsensingpampportalLM7301Blog Post: Go differential to differentiate your precision designhttp://e2e.ti.com/blogs_/b/analogwire/archive/2016/08/11/go-differential-to-differentiate-your-precision-designThu, 11 Aug 2016 15:50:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:d3e84ff8-78bf-440a-8412-d6ee5405e2a0Jason ClarkWhile more of the industry’s newest high-resolution, precision analog-to-digital converters (ADCs) implement differential inputs to maximize performance, many designers still choose to use single-ended amplifiers because that’s what they’re comfortable with. But low-power fully differential amplifiers (FDAs) offer many system advantages without sacrificing precision. In this post, I will use the new THS4551 low noise, precision, 150MHz FDA as an example of how to realize many of the benefits of fully differential amplifiers. FDAs enable simple single-ended to differential signal conversion with direct current (DC) coupling. In Figure 1, you can see three different examples of driving a single-ended signal into the differential input of an ADC. However, the FDA offers lower power, lower noise and an improved dynamic range. (a) (b) (c) Figure 1: Pseudo differential input (a); dual operational amplifier (op amp) method (b); and fully differential amplifier method (c) The FDA architecture can help significantly lower total harmonic distortion (THD) by reducing HD2. Implementing the FDA method shown in Figure 1c, you can achieve an improvement of >4dB in THD. This >4dB improvement can result in an overall system performance improvement, or give you the flexibility to use a lower-power/lower-bandwidth amplifier to meet the same THD. A single FDA (Figure 1c) will have 1/√2 lower noise for the same power than a pair of single-ended op amps (Figure 1b). For example, an op amp with an input-voltage noise of 3nV√Hz will have a total input-voltage noise of 3*√2 nV/√Hz in the dual op amp circuit shown in Figure 1b. The FDA can operate from a single supply voltage and still accept bipolar input signals. Figure 2 shows the THS4551 accepting a 20Vpp input (0V common mode) and outputting 8Vpp with a 2.5V common mode. This capability allows you to reduce system complexity by eliminating the negative power supply and any unnecessary signal-attenuation stages. Figure 2: FDA attenuation example FDAs include a common-mode output loop to perfectly match the expected ADC input common mode. The Vocm pin sets the output common mode of the amplifier. You can leave this pin floating if your desired common mode is at the midpoint of the supplies. As shown in Figure 3, TI FDAs are offered in a number of small packages including 2mm-by-2mm QFN, making them suitable for use in even the most space-constrained application. (a) (b) (c) Figure 3: 2mm-by-2mm 10-pin QFN (a); 3mm-by-3mm 16-pin QFN (b) ; 5mm-by-3mm 8-pin VSSOP (c) TI’s new THS4551 is one of the highest-precision FDAs in the industry, with ±0.175mV input offset and <2V/C offset drift. This enables the improved system performance and minimizes the need for costly and time-consuming system calibrations. The evaluation module (EVM) for TI’s new ADS127L01 24-bit 512Ksps delta-sigma ADC offers an example of the power of FDAs. The EVM for the ADC implements an ADC driver using the THS4551 configured as a multifeedback (MFB) filter. As Figure 4 shows, the ADC-plus-driver pair achieves a signal-to-noise ratio (SNR) of 110.6dB and a THD of 119.1dB with a 1kHz input signal. As shown in the ADS127L01 data sheet (and the goal of all ADC driver implementations), the performance of the THS4551 does not have any impact on the performance of the data converter. This level of performance was achieved while adding less than 7mW of system power, making the THS4551 an essential part of designs that require the lowest power while also delivering the best harmonic distortion and precision. Figure 4: ADS127L01 with THS4551 spectrum If your ADC has a differential input, a precision FDA, such as the THS4551 could be a good choice to simplify your system design and enable low noise, low power, and low harmonic distortion. What is your experience designing with FDAs? Login and leave a comment below about your experience. Additional resources Try an FDA in a system design using one of our TI Designs reference designs, including: Data Acquisition Optimized for Lowest Distortion, Lowest Noise, 18 bit, 1Msps Reference Design (TIPD115). Ultrasonic Water Flow Measurement Reference Design (TIDM-ULTRASONIC-WATER-FLOW-MEASUREMENT) . High Performance Single Ended to Differential Active Interface for High Speed ADC Developed by Dallas Logic Corp. (TIDA-00294). Read more blogs on differentiated amplifiers . Read application notes on FDAs: Using single-supply fully diff. amps with neg. input voltages to drive ADCs Using fully differential op amps as attenuators Analysis of fully differential amplifiers Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalFully Differential Amplifiershsamps_fdaTHS4551TIDA-00294TIPD115smart gridtest measurementTIDM-LC-WATERMTRpampportalADS127L01Precision Op AmpsTIDM-ULTRASONIC-WATER-FLOW-MEASUREMENTfactory automationFully differential amp blogsBlog Post: How to design cost-sensitive battery-monitoring circuitshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/07/15/how-to-design-cost-sensitive-battery-monitoring-circuitsFri, 15 Jul 2016 13:01:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:2bd1a109-0334-46a1-bd51-4cb93a035816Collin WellsIn portable electronics designs, typical battery-monitoring systems measure battery voltage and battery current to detect when the battery needs charging or replacement. In this post, I’ll demonstrate battery-voltage and current-monitoring circuitry for cost-optimized systems using operational amplifiers (op amps). Op amps used in battery-monitoring circuitry must meet the required accuracy levels while consuming minimal power-supply quiescent current, i Q­ , to conserve battery life. Table 1 lists the key specifications for two new op amps, the TLVx369 and TLVx379 families, which are designed for low-power, cost-sensitive applications. Table 1 : Key specifications for TLV369 and TLV379 op amps In figure 1, you will see an example battery-voltage measurement circuit using the TLV379 configured as a unity-gain buffer. To prevent violations of the amplifier’s common-mode input voltage range or output voltage swing, the battery is divided down using R1 and R2. In this case, a 1.8V-5.5V battery voltage will create a 0.393V-1.2V output voltage, which fits within the common 0V-1.2V range for analog-to-digital converters (ADCs) on many low-power microcontrollers. Equation 1 shows the transfer function for the circuit in Figure 1. Be sure to use high-value resistors for the divider to minimize current consumption. You can place a low-pass filter at the output of the circuit to limit the signal bandwidth and output noise. However, like most low-power op amps , the TLV379 does not perform well while driving capacitive loads, so check the stability of the circuit when designing output filters with capacitances to GND. Figure 1: Battery-voltage measurement circuit using the TLV379 The main error contributors in the circuit shown in Figure 1 are the tolerance of the resistors in the divider and the offset voltage of the op amp. Other error sources come from the op-amp’s CMRR and the input bias current flowing through the voltage-divider resistors. Table 2 uses the typical specifications for the TLV379 listed in Table 1 to calculate expected circuit performance. The resistor divider tolerance, R TOL , is set to 0.1%. Table 2 : Error calculations for the TLV379 voltage-measurement circuit shown in Figure 1 Measuring the voltage drop across a low-side current-shunt resistor is often the simplest method to determine battery/load current. Figure 2 shows an example low-side current-sensing circuit using the TLV379 . The circuit in Figure 2 was designed to create a 0V-1.2V output voltage for a 0A-1A load current, i LOAD . Equations 2 and 3 calculate the input voltage, V IN­ , and output voltage, V OUT , respectively; you can use these equations to adjust the circuit for other ranges. In Figure 2, the shunt voltage, V SHUNT , was limited to 100mV with the maximum 1A load current; you could use other V SHUNT values depending on what the load can tolerate. Be sure to make a good Kelvin (or four-wire) connection across the shunt resistor, R S , to reduce the effects of printed circuit board (PCB) impedances. Figure 2: Low-side battery current-measurement circuit using the TLV379 Table 3 lists the error calculations for the circuit in Figure 2. Shunt-resistor values with 100dB of CMRR over the full supply voltage range. The circuit was designed to keep the output voltage below 1.2V for a 1A load current and 5.5V battery voltage. Equations 4, 5 and 6 show simplified transfer functions for the circuit. The bias voltage, V BIAS , created from the resistor divider pushes the output voltage away from the negative rail, allowing for current measurements down to 0A. You must combine this circuit with the battery-voltage measurement to obtain the value of V BIAS . While I didn’t include them in this post, the error calculations for the circuit in Figure 3 are similar to those shown in Table 3. Figure 3: High-side battery-current measurement circuit using the TLV369 Battery-powered electronics almost always include battery-voltage and current-monitoring circuitry for gauging and protection purposes. In this post, I provided circuit examples for a voltage-measurement circuit and both low- and high-side current-sensing circuits using some of our newest op amps, which provide an excellent price-to-performance ratio for cost-sensitive systems. Have questions about other op-amp designs? Log in and leave a comment. Additional resources Read my colleague, Pete Semig’s blog, “ Resistor Divider Drift: 5ppm + 5ppm = 5ppm ” which I used to determine the gain error from the resistor divider in circuit 1. See TI’s portfolio of performance op amps for cost-conscious applications . Watch more than 40 on-demand precision amplifier training videos in our TI Precision Labs – Op Amps series. Find commonly used analog design formulas in the Analog Engineer’s Pocket Reference e-book.opampportalbattery monitorgeneral purpose op amp blogsprecision op amp blogsgeneral-purpose op ampsUltra-low-power op amp blogspampportalPrecision Op Ampscurrent sensingTLV blog seriesTLV379TLV369Blog Post: Designing high-performance, cost-sensitive transimpedance op-amp circuitshttp://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/06/20/how-to-get-the-best-performance-from-your-transimpedance-designsMon, 20 Jun 2016 08:10:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:88210897-087e-4e4d-b837-63d9187ec105Collin WellsThis post is co-authored by Raphael Puzio . Photodiode-based light sensing is a common application of operational amplifiers (op amps) used in medical equipment, industrial automation, robotics, point-of-sale machines, drones, smoke detectors and building automation equipment. This blog demonstrates how to build a cost-sensitive, accurate photodiode circuit. A photodiode sensor produces a current proportional to the light level presented to it. Depending on the application, the photodiode is operated in either a photovoltaic or photoconductive mode; each has its own merits, which Bruce Trump discussed in detail in this post from his blog, The Signal . Most applications operate the photodiode in photoconductive mode, with an op amp in a transimpedance configuration to amplify the current. In photoconductive mode, the photodiode is held at a zero-volt (Figure 1a) or reverse voltage bias (Figure 1b), preventing it from forward biasing. Figure 1: Photodiode in photoconductive mode with zero-volt bias (a); or reverse-voltage bias (b) Equation 1 calculates the direct current (DC) transfer function for the circuits shown in Figure 1 (note that the photodiode current (i D ) is flowing away from the op-amp inverting node): The three-step process outlined in John Caldwell’s series on transimpedance amplifiers (see part 3, “ What op amp bandwidth do I need ?”) determines the minimum required op-amp gain bandwidth for transimpedance configurations. The minimum bandwidth is based on the required transimpedance gain and signal bandwidth, along with the total capacitance presented to the inverting node of the op amp. The diode capacitance often dominates the inverting-node capacitance, but don’t forget to include the effects of the op-amp input capacitance. We summarized the three steps explained in John’s posts here for quick reference. 1. Choose the maximum feedback capacitance (C F ) based on the feedback resistor (R F ) and the signal -3dB bandwidth (f P ) (Equation 2): 2. Calculate the total capacitance (C IN ) at the inverting input of the amplifier. For the circuits shown in Figure 1, this is equal to Equation 3: where C J is the diode junction capacitance, C D is the op-amp differential input capacitance and C CM2 is the op-amp inverting input common-mode input capacitance. 3. Calculate the minimum required op-amp gain bandwidth product (f GBW ) (Equation 4): By following these three simple steps, you can avoid many of the stability and performance issues commonly associated with transimpedance amplifier circuits by selecting an amplifier with sufficient bandwidth to perform the required transimpedance gain at the desired signal bandwidth. Along with meeting bandwidth requirements, the op amp must also meet the system’s DC accuracy requirements. The most important DC specification in many transimpedance applications is the input bias current (i B ) of the op amp. i B will directly sum or subtract with the input signal current, which can cause large errors depending on the magnitude of i B compared to the signal current. The example shown in Figure 2 uses a 5MΩ resistor to apply a 5MV/A gain to a 100nA full-scale input current. With the input bias current set to 0A, the full-scale output voltage is 500mV – which is expected based on the transfer function in Equation 1. The circuit on the right in Figure 2 displays the effects of the same circuit with an op amp i B of 10nA. In this case, the output voltage is 450mV, which shows that the 10nA input bias current caused a 50mV (or 10%) error from the ideal 500mV output signal. Figure 2: Input bias current effects in transimpedance amplifier circuits Equation 5 calculates the percentage error of the full-scale range (%FSR) based on the full-scale input current (i IN_FS ) and the op amp’s i B : The TLV600x devices are a new family of high-performance general-purpose amplifiers for a wide variety of cost-conscious transimpedance applications, such as consumer drones, POS machines, smoke detectors and building automation equipment. The key features that make the TLV600x great for transimpedance applications is the wide bandwidth of 1MHz, low typical input bias currents of 1pA and a low input capacitance of 6pF total. Other beneficial features are a low quiescent current of 100µA maximum, rail-to-rail input and output swings, and low current and voltage noise densities of 5fA/√Hz and 28nV/√Hz, respectively. Table 1 lists different transimpedance gain and bandwidth combinations for the TLV600x based on Equations 1 through 4. Be sure to keep the total input capacitance below the maximum input capacitance (C IN­_MAX ) to avoid stability issues. Note: f GBW is 1MHz, C D is 1pF and C CM is 5pF for the TLV600x. Table 1 : Quick design calculator for TLV600x transimpedance applications Figure 3 shows the simulated step-response results for a 1MV/A gain and 50kHz bandwidth with the maximum 54pF of input capacitance from the photodiode. The output overshoot and ringing are minimal, indicating a stable design. Figure 3: TLV6000 step-response results; gain = 1MV/A, bandwidth = 50kHz Many applications use op amps in the transimpedance configuration to amplify low-level currents. Designing the transimpedance circuit can be simplified to a few easy steps. First, follow the three steps from John’s blog posts to select the required op-amp bandwidth. Then sort the remaining results to find a device with an i B specification that meets the system’s DC requirements. Do you have questions about transimpedance op-amp designs? Log in and leave a comment below letting us know your experience with transimpedance configurations or any questions you have. Additional resources Find commonly used analog design formulas in the “ Analog Engineer’s Pocket Reference ” e-book. Read the other two parts in John Caldwell’s three-part blog series on transimpedance circuits: “ Part I – What op amp bandwidth do I need? (Transimpedance Amplifiers) .” “ Part II – What op amp bandwidth do I need? (Transimpedance Amplifiers) . See TI’s portfolio of performance op amps for cost-conscious applications .opampportalgeneral purpose op amp blogsgeneral-purpose op ampspampportalPrecision AmplifierPrecision Op AmpsTLV blog seriesBlog Post: Amp up your cans: pop/click suppression in headphone amplifiers (Part 5)http://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/04/15/amp-up-your-cans-pop-click-suppression-in-headphone-amplifiers-part-5Fri, 15 Apr 2016 09:45:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:2e6ed2cf-7955-446b-b046-948dfc57d2edBharath VasanThis post was co-authored with John Caldwell . In the final installment of this five-part blog series, I will discuss pop/click noise in operational amplifiers (op amps) driving headphone loads, and some techniques to minimize them. The previous posts discussed power considerations for headphone loads, impedance of headphones, and sources of headphone amplifier stability and distortion. Undesirable audible noise resulting from applying power or changing operational modes in an audio system is commonly called a pop or a click. This effect is of extreme concern in high-fidelity headphone systems due to the high efficiency of the headphone drivers. Even relatively small transients in the signal voltages will produce loud and unpleasant sounds from the headphones. In order to improve the user experience and prevent damage to headphones or other sensitive electronics, many systems employ circuitry to suppress transient signals from the amplifier output during power-up. Figure 1: Op amp driving a headphone load Let’s look at some of the common sources of pop/click noise in an op amp. Figure 1 is a simplified block diagram of an op amp driving a headphone load. It consists of two stages – a gain stage and an output buffer – as well as a compensation capacitor, Cc, and an offset voltage, Vos. The output buffer is a unity gain stage capable of sinking and sourcing current as shown. Pop/click noise is typically associated with transients at the output pin of the op amp. The chief sources of pop/click noise are: Power-supply ramp : During power-on, VCC and VEE gradually ramp up. Until the op amp reaches its minimum supply requirement, it has not yet attained steady-state operation and is not regulating the output. During this period, large transients could result at the output pin. A power-off where VCC and VEE ramp down could be accompanied by similar transients, as could asymmetric ramping of VCC and VEE. Enabling/disabling the amplifier : Many op amps include options to enable or shut down the device. The enable command turns on the op amp’s internal bias circuitry. As the bias currents come up, the gain stage starts charging the compensation capacitor to the appropriate voltage, the sourcing and sinking currents in the output buffer start stabilizing, and the op amp approaches steady state. During this phase, unequal sourcing and sinking currents in the output buffer result in current being injected into the load. This could result in significant pop/click noise. Similar transients can occur after a shutdown command. Offset voltage of the amplifier : An interesting source of pop/click noise is the offset voltage. Once the amplifier reaches steady state, the output of the amplifier jumps to the offset (or Vos/β, β = feedback factor) of the amplifier. If the offset is large enough, the change in the output is audible. Amplifiers with integrated pop-click suppression can significantly simplify audio systems. The OPA1622 high-performance headphone amplifier includes a solution to suppress pop/click noise when the part is enabled or in shutdown mode. Figure 2 shows a simplified block diagram of the OPA1622 . Figure 2: OPA1622 block diagram The enable circuitry (ENC) maintains control of the input and output stage when the amplifier is enabled or in shutdown mode. When enabled, the ENC steadily transitions to power the gain stage and the output stage. Upon shutdown, the ENC controls the charge on the compensation capacitor and disables the gain stage and the output stage. The carefully optimized transition into and out of shutdown ensures that very small currents are injected into the headphone load, thus minimizing pop/click noise. During the OPA1622 ’s development, we focused on optimizing the offset voltage and its contribution to pop/click noise. The OPA1622 has an offset voltage of 50µV typical and 500µV max. Figures 3 and 4 show the performance of the OPA1622 during enable and disable, respectively. Only very small output transients accompany the enable or disable events. The output transients occur over a very short duration and are therefore unlikely to be audible in headphone applications. Figure 3: Output voltage when enable transitions high (32Ω) Figure 4: Output voltage when enable transitions low (32 Ω ) Several factors can affect pop/click noise performance in op amps driving headphones. By including a pop/click noise-suppression circuit, the OPA1622 can improve the overall user experience in audio applications. Do you have experience with pops and clicks in your system? Log in, leave a comment and let us know about your system design. Additional resources Watch a four-part TI Precision Labs – Op Amps training course on low-distortion design . Search TI Designs – Precision reference designs to help get your next design started. Download the OPA1622 TINA-TI™ SPICE macromodel . Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalamp up your cans blogaudioopampportalPrecision amplifier blogpampportalOPA1622Precision Op Ampstina-tiBlog Post: Where did the op amp ground pin go?http://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/03/18/where-did-the-ground-pin-goFri, 18 Mar 2016 08:30:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:5518d858-9fb5-4cee-9f77-89138150e0a3Soufiane BendaoudDid you know that there are operational amplifiers (op amps) with a ground pin? If you’re an experienced analog engineer, you probably know that, but those who haven’t worked much with older op amps might be asking themselves, “What’s a ground pin?” Integrated circuit (IC) manufacturers used to label the ground pin, which is supposed to indicate zero volts, “GND.” Today, manufacturers like TI label the pin on new op amps “V-,” or “N/C” (negative supply and no connect). This is because GND is supposed to indicate zero volts, but an op amp never puts out exactly zero volts, making the original label confusing. The LMV321 , LMV324 and LMV358 , which are more than 20 years old, are good examples of amplifiers that still use the GND label. You will see in Figure 1, the LMV358 and LMV321 pinouts, including the GND label. Figure 1: Top view of the LMV358 and LMV321 pinout showing the ground pin The LM710 comparator is another example of an older device that is still very relevant for today’s systems and uses the GND label. However, in this instance, it actually indicates where ground, or zero volts is, as you will see in figure 2. Figure 2: LM710 pinout with the GND labeled Besides the rare examples, like the ones I mentioned above, it seems the ground pin label has disappeared altogether, at least in op amps. The old GND label gave way to appellations such as “single supply,” which can also be misleading. Does “single supply” mean you cannot use the amplifier in a split-supply fashion? Of course you can, as long as you don’t violate your device’s common-mode input range. But many engineers who are not accustomed to designing with older op amps might not know this. A shortcoming of the single-supply op amp is that it can only accept positive voltages (at most a few hundred millivolts below ground) and produce positive voltages at its output. The “single supply” label was meant for devices targeted at high-volume, low-cost consumer applications. After all, an additional negative power supply adds additional cost. Many options exist to get a true zero output without worrying whether there is a pin labeled GND. For a dual-supply design, consider switching regulators with dual outputs, such as the TPS65133 or the TPS65130 , which will synthesize a dual supply. If your design uses just one positive power supply, consider a more cost-effective solution with the LM7705 , one of my favorite true-zero-output generators. All three of these devices will enable you to get a true zero at the output of the amplifier. Yet another option is to use the TLE2446 single supply with a rail splitter, which allows you synthesize a reference (mid-supply). I would like to hear your comments about your experience with the GND pin. Log in and drop me a line below in the comments. Let me know why you think it was there in the first place. Additional resources Learn more about amplifiers by downloading the “ Best of Baker’s Best Amplifiers ” e-book by my colleague and EDN columnist Bonnie Baker. Read a blog on the basics of how to layout a PCB for an op amp by Soufiane’s colleague Timothy Claycomb. Visit the Precision Amplifier forum to learn more from other engineers. Download the “ Analog Engineer’s Pocket Reference ” e-book. Learn about TI’s entire portfolio of amplifier ICs and find technical resources.opampportalTPS65130LMV321LM71LMV358lmv324LM7705pampportalPrecision op amp blogTPS65133Blog Post: Amp up your cans: Distortion in headphone amplifiers (Part 4)http://e2e.ti.com/blogs_/archives/b/precisionhub/archive/2016/03/04/amp-up-your-cans-part-3b-distortion-in-headphone-amplifiersFri, 04 Mar 2016 07:00:00 GMTcb01d8b2-d089-468d-babb-77d1d8683490:a2d99f5a-eb8f-4871-808c-eda095733251John CaldwellThis post is co-authored by Bharath Vasan . In part three of this five-part blog series , we introduced a simple control-loop model for distortion in an operational amplifier (op amp), repeated here in Figure 1. We also included a pop quiz that we’ll continue discussing in this post, so please read that installment first if you haven’t done so already. Everything we discuss here will make more sense that way. The model in Figure 1 is useful in illustrating how a source of distortion would be attenuated by the loop gain of the system, but it does not reveal anything about the actual mechanism of distortion and why it would increase with low load impedances. Figure 1: A control-loop diagram of an op amp that includes an internal source of distortion and the resulting transfer function In an op amp’s output stage, the distortion does not come from a separate source injected into the loop; it arises from variations in the impedance of the output stage with the load voltage or current. As illustrated in Figure 2, the actual gain of the output stage (A OS ) can be represented as a voltage divider consisting of the load impedance (R L ) and the impedance of the output stage (R OUT ). Figure 2: The output stage gain depends on output impedance and load impedance Therefore, the gain of the output stage can never be exactly 1, and it also depends on the load impedance (Equation 1): R OUT separates into two terms: a static impedance (R O ) and a dynamic one that varies as a function of the output voltage and output current (ΔR[V O , I O ]). See Equation 2: It is this varying output impedance that produces distortion. For example, if R OUT decreases when the output stage is sourcing current, then the overall gain of the output stage would be greater for positive voltages compared to negative ones, and would distort the input signal. Substituting the expression for R OUT back into Equation 1 offers some insight into the role that load impedance plays: Notice that ΔR O is divided by the load impedance. Therefore, reducing the load impedance will cause greater variations in the gain of the output stage and produce more distortion (answer C from the pop quiz in part 3 of this series ). To show this effect, my colleague Bharath Vasan simulated the output stage gain versus the output voltage for the OPA1688 , a general purpose audio op amp, and the OPA1622 , an audio op amp designed for low impedance loads. He used 604-Ω and 32.4-Ω loads for these simulations. Figures 3 and 4 show the results. In the 604-Ω load case, both amplifiers displayed stable gain over the range of output voltages simulated. But in the 32.4-Ω case, the differences between the two output stages became apparent. The gain deviation was worst near 0V, where the load current transitioned from one output transistor to another. This is commonly known as output crossover distortion. In the OPA1622 , the output crossover region is greatly reduced, and the gain deviation in this region is minimized. Figure 3: Output stage gain vs. output voltage for a 604-Ω load Figure 4: Output stage gain vs. output voltage for a 32.4-Ω load Congratulations to those of you who answered “D” to the pop quiz in part 3 ! High-frequency distortion depends on both the loop gain of the amplifier and the distortion of the output stage. When developing the OPA1622 , we took a dual approach to reduce distortion. Not only did we maximize the open-loop gain of the amplifier at audio frequencies, but we reduced the intrinsic distortion of the output stage by addressing the dynamic component of the output impedance. The result of this improved output stage is instantly recognizable in the distortion measurements. The measurement performed in Part 3 on the OPA1688 is now repeated in Figure 5 with the OPA1622 . Notice the improvement in high frequency distortion for all load impedances. Figure 5: THD+N vs. frequency curve for an OPA1622 with various loads In the final installment of this blog series , we’ll take a look at the shutdown feature of the OPA1622 and how we minimized audible “clicks” and “pops” when the amplifier is enabled and disabled. Additional resources Watch a four-part TI Precision Labs – Op Amps training course on low-distortion design . Search TI Designs – Precision reference designs to help get your next design started. Read the OPA1622 and the OPA1688 datasheets.opampportalamp up your cans blogsOPA1688audioopampportalPrecision amplifier blogpampportalOPA1622Precision Op AmpsAudio op amp blog