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Power transformer magnetizing inductance and current (UCC28950)

Florin Ciortea
Prodigy 90 points
Other Parts Discussed in Thread: UCC28950

Hi,


The primary magnetizing inductance for the power transformer is calculated in UCC28950 application note in eq. 9 page 3. I presume the 0.5 term within the denominator comes from half the switching frequency of the output inductor (the operational frequency of the transformer) and is better linked to fs rather than the output ripple inductor current - please correct me if wrong.

In eq. 17 page 5 the primary magnetizing current is determined based on output inductor frequency. Is this ok? I believe that we have to consider the transformer operatig frequency as it is a transformer parameter. Is it possible to find how the formula is derived (both: eq. 9 and 17)?

Also this parameter is used to set the sensing network for the current transformer, so is quite important to know the correct value. I believe the IP1 term in eq. 85 page 15 is similar to IPP term in 18 page 5 - please confirm!


UCC28950 application note link: http://www.ti.com/lit/an/slua560c/slua560c.pdf

Thank you!

  • 0
    TI__Mastermind 31580 points
    Hi Florin,

    I have forwarded the post to my colleague who will answer this tomorrow.

    Regards

    Peter
  • 0 in reply to Peter Meaney
    TI__Mastermind 25570 points

    Hi Florin

    This inequality (eq 9) is used to ensure that the contribution to the slope of the Current Sense signal due to magnetizing current was not more than half that due to the inductor ripple current down-slope, reflected to the primary of the transformer, this is the origin of the 0.5 term in the expression. The Vin term is the voltage applied to the magnetizing inductance - the rest of the expression reduces to a calculation of the recriprocal (dt/dI) of the output inductor current down-slope during the off time reflected through the transformer turns ratio. The off time is defined by the (1-D) term along with the switching frequency as seen by the output inductor which of course is twice the switching frequency seen on the primary. In this equation, fs is the switching frequency seen by the inductor ie. 200kHz.

    The 0.5 term is there to ensure that the current slope condition mentioned earlier is met.

    Of course L = V dt/dI and we use this expression to limit Lmag > Vin dt/dI. (1-Dtyp) was chosen simply because it reflects normal operating conditions - it is not a worst case condition.

    Eq17, the ΔILmag of course depends on the amount of time for which the input voltage is applied to the transformer winding. On any given switching cycle this Δ happens twice - once when the AD pair are turned on and once when the BC pair are turned on - I'm using the switch designators in the same way as in the data sheet. to this Δ happens twice each switching cycle - ie at fs = 200kHz as described in the app note. Eq 17 calculates the ΔI. This is then added to the IPP value calculated in Eq 18 to give the total RMS current (Eq 19)

    You are correct - The IP1 term in eq. 85 page 15 is similar (actually the same, just rewritten slightly) to IPP term in 18 page 5

    Hopefully this answers your question - Please feel free to re-post any further questions

    Regards

    Colin

  • 0 in reply to Colin Gillmor
    Prodigy 90 points

    Hi Colin,


    Many thanks for your competent and comprehensive answer! I really appreciate!

    Still if possible I have some additional question to ask. I have designed a phase shifted ZVS converter and implemented it practically. It is intended to charge LiFePO4 cells within a battery management system. The input voltage ranges between 28V and 36V (I believe this is not an issue although is not an off-line application with higher input) with 30A at 3.75V output (100kHz on output inductor). I have designed the magnetics by myself and as I am working at low input voltages I relied solely on the leakage inductance to achieve resonant transitions. I have two big questions to pose: 

    1. I closed the control loop with a type 2 error amplifier as the double pole due to the output inductance is shifted out in frequency by the current mode control at 1/4 the switching frequency. I believe I have closed the loop quite ok (3kHz crossover at 87deg flat, theoretically). However, the converter is strongly oscillating from 35V down. At 36V the voltage on the main transformer is trembling a bit due to control loop and this is not ok either I presume (I believe the duty cycle for a fixed input voltage should be crisper). The real behavior is very similar with the simulated one. As a result I tried to tune the control loop in several ways but no result. Do you have any idea what I have missed. The output inductor current has a strange wave which clusters the every other third teeth in some kind of hump. The attached file contains a simulation plot.

    2. I had a look on the MOSFET transitions. For the sluggish leg the transition is not really resonant so I will add a shim in order to catch resonance. On the other hand for the leading leg although the Vds drops before Vgs rise, the rising front of Vgs is not steep at all. Actually the Vgs ramps up in a bumpy manner. Are you familiar which such a behavior. Is this ok or I should check for some design flaws. I attached some plots fr a better understanding.

    Many thanks for your support! Wish you all the best!https://e2e.ti.com/cfs-file/__key/communityserver-discussions-components-files/196/Plots-and-sim.7z